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functional block diagram pga buffer charge balancing a/d converter sigma-delta modulator digital filter ref in(C) ref in(+) av dd dv dd a = 1C128 mclk in mclk out reset ain(+) ain(C) serial interface register bank sclk cs din dout drdy clock generation agnd dgnd AD7715 rev. b information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by analog devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. no license is granted by implication or otherwise under any patent or patent rights of analog devices. a 3 v/5 v, 450 m a 16-bit, sigma-delta adc AD7715* features charge-balancing adc 16 bits no missing codes 0.0015% nonlinearity programmable gain front end gains of 1, 2, 32 and 128 differential input capability three-wire serial interface spi?, qspi?, microwire? and dsp compatible ability to buffer the analog input 3 v (AD7715-3) or 5 v (AD7715-5) operation low supply current: 450 m a max @ 3 v supplies low-pass filter with programmable output update 16-lead soic/dip/tssop cmos construction ensures very low power dissipation, and the power-down mode reduces the standby power consumption to 50 m w typ. the part is available in a 16-lead, 0.3 inch-wide, plastic dual-in-line package (dip) as well as a 16-lead 0.3 inch- wide small outline (soic) package and a 16-lead tssop package. product highlights 1. the AD7715 consumes less than 450 m a in total supply current at 3 v supplies and 1 mhz master clock, making it ideal for use in low-power systems. standby current is less than 10 m a. 2. the programmable gain input allows the AD7715 to accept input signals directly from a strain gage or transducer remov- ing a considerable amount of signal conditioning. 3. the AD7715 is ideal for microcontroller or dsp processor applications with a three-wire serial interface reducing the number of interconnect lines and reducing the number of opto-couplers required in isolated systems. the part con- tains on-chip registers which allow software control over output update rate, input gain, signal polarity and calibration modes. 4. the part features excellent static performance specifications with 16-bits no missing codes, 0.0015% accuracy and low rms noise (<550 nv). endpoint errors and the effects of temperature drift are eliminated by on-chip calibration op- tions, which remove zero-scale and full-scale errors. general description the AD7715 is a complete analog front end for low frequency measurement applications. the part can accept low level input signals directly from a transducer and outputs a serial digital word. it employs a sigma-delta conversion technique to realize up to 16 bits of no missing codes performance. the input signal is applied to a proprietary programmable gain front end based around an analog modulator. the modulator output is pro- cessed by an on-chip digital filter. the first notch of this digital filter can be programmed via the on-chip control register allow- ing adjustment of the filter cutoff and output update rate. the AD7715 features a dif ferential analog input as well as a dif- ferential reference input. it operates from a single supply (+3 v or +5 v). it can handle unipolar input signal ranges of 0 mv to +20 mv, 0 mv to +80 mv, 0 v to +1.25 v and 0 v to +2.5 v. it can also handle bipolar input signal ranges of 20 mv, 80 mv, 1.25 v and 2.5 v. these bipolar ranges are referenced to the negative input of the differential analog input. the AD7715 thus performs all signal conditioning and conversion for a single- channel system. the AD7715 is ideal for use in smart, microcontroller or dsp based systems. it features a serial interface that can be config- ured for three-wire operation. gain settings, signal polarity and update rate selection can be configured in software using the input serial port. the part contains self-calibration and system calibration options to eliminate gain and offset errors on the part itself or in the system. spi and qspi are trademarks of motorola, inc. microwire is a trademark of national semiconductor corporation. * protected by u.s. patent no: 5,134,401. see page 30 for data sheet index. one technology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. tel: 781/329-4700 world wide web site: http://www.analog.com fax: 781/326-8703 ? analog devices, inc., 1998
parameter a version 1 units conditions/comments static performance no missing codes 16 bits min guaranteed by design. filter notch 60 hz output noise see tables v to viii depends on filter cutoffs and selected gain integral nonlinearity 0.0015 % of fsr max filter notch 60 hz unipolar offset error see note 2 unipolar offset drift 3 0.5 m v/ c typ bipolar zero error see note 2 bipolar zero drift 3 0.5 m v/ c typ positive full-scale error 4 see note 2 full-scale drift 3, 5 0.5 m v/ c typ gain error 6 see note 2 gain drift 3, 7 0.5 ppm of fsr/ c typ bipolar negative full-scale error 2 0.0015 % of fsr max typically 0.0004% bipolar negative full-scale drift 3 1 m v/ c typ for gains of 1 and 2 0.6 m v/ c typ for gains of 32 and 128 analog inputs/reference inputs specifications for ain and ref in unless noted input common-mode rejection (cmr) 90 db min at dc. typically 102 db normal-mode 50 hz rejection 8 98 db min for filter notches of 25 hz, 50 hz, 0.02 f notch normal-mode 60 hz rejection 8 98 db min for filter notches of 20 hz, 60 hz, 0.02 f notch common-mode 50 hz rejection 8 150 db min for filter notches of 25 hz, 50 hz, 0.02 f notch common-mode 60 hz rejection 8 150 db min for filter notches of 20 hz, 60 hz, 0.02 f notch common-mode voltage range 9 agnd to av dd v min to v max ain for buf bit of setup register = 0 and ref in absolute ain/ref in voltage 8 agnd C 30 mv v min ain for buf bit of setup register = 0 and ref in av dd + 30 mv v max absolute/common-mode ain voltage 9 agnd + 50 mv v min buf bit of setup register = 1 av dd C 1.5 v v max ain dc input current 8 1 na max ain sampling capacitance 8 10 pf max ain differential voltage range 10 0 to +v ref /gain 11 nom unipolar input range (b/u bit of setup register = 1) v ref /gain nom bipolar input range (b/u bit of setup register = 0) ain input sampling rate, f s gain f clk in /64 for gains of 1 and 2 f clk in /8 for gains of 32 and 128 ref in(+) C ref in(C) voltage +2.5 v nom 1% for specified performance. functional with lower v ref ref in input sampling rate, f s f clk in /64 logic inputs input current 10 m a max all inputs except mclk in v inl , input low voltage 0.8 v max dv dd = +5 v v inl , input low voltage 0.4 v max dv dd = +3.3 v v inh , input high voltage 2.4 v min dv dd = +5 v v inh , input high voltage 2.0 v min mclk in only v inl , input low voltage 0.8 v max dv dd = +5 v v inl , input low voltage 0.4 v max dv dd = +3.3 v v inh , input high voltage 3.5 v min dv dd = +5 v v inh , input high voltage 2.5 v min dv dd = +3.3 v logic outputs (including mclk out) v ol , output low voltage 0.4 v max i sink = 800 m a except for mclk out 12 . dv dd = +5 v v ol , output low voltage 0.4 v max i sink = 100 m a except for mclk out 12 . dv dd = +3.3 v v oh , output high voltage 4.0 v min i source = 200 m a except for mclk out 12 . dv dd = +5 v v oh , output high voltage dv dd C 0.6 v v min i source = 100 m a except for mclk out 12 . dv dd = +3.3 v floating state leakage current 10 m a max floating state output capacitance 13 9 pf typ data output coding binary unipolar mode offset binary bipolar mode AD7715-5Cspecifications (av dd = +5 v, dv dd = +3 v or +5 v, ref in(+) = +2.5 v; ref in(C) = agnd; f clk in = 2.4576 mhz unless otherwise noted. all specifications t min to t max unless otherwise noted.) rev. b C2C
parameter a version 1 units conditions/comments static performance no missing codes 16 bits min guaranteed by design. filter notch 60 hz output noise see tables ix to xii depends on filter cutoffs and selected gain integral nonlinearity 0.0015 % of fsr max filter notch 60 hz unipolar offset error see note 2 unipolar offset drift 3 0.2 m v/ c typ bipolar zero error see note 2 bipolar zero drift 3 0.2 m v/ c typ positive full-scale error 4 see note 2 full-scale drift 3, 5 0.2 m v/ c typ gain error 6 see note 2 gain drift 3, 7 0.2 ppm of fsr/ c typ bipolar negative full-scale error 2 0.003 % of fsr max typically 0.0004% bipolar negative full-scale drift 3 1 m v/ c typ for gains of 1 and 2 0.6 m v/ c typ for gains of 32 and 128 analog inputs/reference inputs specifications for ain and ref in unless noted input common-mode rejection (cmr) 90 db min at dc. typically 102 db normal-mode 50 hz rejection 8 98 db min for filter notches of 25 hz, 50 hz, 0.02 f notch normal-mode 60 hz rejection 8 98 db min for filter notches of 20 hz, 60 hz, 0.02 f notch common-mode 50 hz rejection 8 150 db min for filter notches of 25 hz, 50 hz, 0.02 f notch common-mode 60 hz rejection 8 150 db min for filter notches of 20 hz, 60 hz, 0.02 f notch common-mode voltage range 9 agnd to av dd v min to v max ain for buf bit of setup register = 0 and ref in absolute ain/ref in voltage 8 agnd C 30 mv v min ain for buf bit of setup register = 0 and ref in av dd + 30 mv v max absolute/common-mode ain voltage 9 agnd + 50 mv v min buf bit of setup register = 1 av dd C 1.5 v v max ain dc input current 8 1 na max ain sampling capacitance 8 10 pf max ain differential voltage range 10 0 to +v ref /gain 11 nom unipolar input range (b/u bit of setup register = 1) v ref /gain nom bipolar input range (b/u bit of setup register = 0) ain input sampling rate, f s gain f clk in /64 for gains of 1 and 2 f clk in /8 for gains of 32 and 128 ref in(+) C ref in(C) voltage +1.25 v nom 1% for specified performance. functional with lower v ref ref in input sampling rate, f s f clk in /64 logic inputs input current 10 m a max all inputs except mclk in v inl , input low voltage 0.8 v max v inh , input high voltage 2.0 v min mclk in only v inl , input low voltage 0.4 v max v inh , input high voltage 2.5 v min logic outputs (including mclk out) v ol , output low voltage 0.4 v max i sink = 100 m a except for mclk out 12 v oh , output high voltage dv dd C 0.6 v min i source = 100 m a except for mclk out 12 floating state leakage current 10 m a max floating state output capacitance 13 9 pf typ data output coding binary unipolar mode offset binary bipolar mode AD7715 AD7715-3Cspecifications (av dd = +3 v, dv dd = +3 v, ref in (+) = +1.25 v; ref in(C) = agnd; f clk in = 2.4576 mhz unless otherwise noted. all specifications t min to t max unless otherwise noted.) C3C rev. b
pa rameter a version units conditions/comments system calibration positive full-scale calibration limit 14 (1.05 v ref )/gain v max gain is the selected pga gain (1, 2, 32 or 128) negative full-scale calibration limit 14 C(1.05 v ref )/gain v max gain is the selected pga gain (1, 2, 32 or 128) offset calibration limit 15 C(1.05 v ref )/gain v max gain is the selected pga gain (1, 2, 32 or 128) input span 15 0.8 v ref /gain v min gain is the selected pga gain (1, 2, 32 or 128) (2.1 v ref )/gain v max gain is the selected pga gain (1, 2, 32 or 128) power requirements power supply voltages av dd voltage (AD7715-3) +3 to +3.6 v for specified performance av dd voltage (AD7715-5) +4.75 to +5.25 v for specified performance dv dd voltage +3 to +5.25 v for specified performance power supply currents av dd current av dd = 3.3 v or 5 v. gain = 1 to 128 (f clk in = 1 mhz) or gain = 1 or 2 (f clk in = 2.4576 mhz) 0.27 ma max typically 0.2 ma. buf bit of setup register = 0 0.6 ma max typically 0.4 ma. buf bit of setup register = 1 av dd = 3.3 v or 5 v. gain = 32 or 128 (f clk in = 2.4576 mhz) 16 0.5 ma max typically 0.3 ma. buf bit of setup register = 0 1.1 ma max typically 0.8 ma. buf bit of setup register = 1 dv dd current 17 digital i/ps = 0 v or dv dd . external mclk in 0.18 ma max typically 0.15 ma. dv dd = 3.3 v. f clk in = 1 mhz 0.4 ma max typically 0.3 ma. dv dd = 5 v. f clk in = 1 mhz 0.5 ma max typically 0.4 ma. dv dd = 3.3 v. f clk in = 2.4576 mhz 0.8 ma max typically 0.6 ma. dv dd = 5 v. f clk in = 2.4576 mhz power supply rejection 18 see note 19 db typ normal-mode power dissipation 17 av dd = dv dd = +3.3 v. digital i/ps = 0 v or dv dd . external mclk in 1.5 mw max buf bit = 0. all gains 1 mhz clock 2.65 mw max buf bit = 1. all gains 1 mhz clock 3.3 mw max buf bit = 0. gain = 32 or 128 @ f clk in = 2.4576 mhz 5.3 mw max buf bit = 1. gain = 32 or 128 @ f clk in = 2.4576 mhz normal-mode power dissipation 17 av dd = dv dd = +5 v. digital i/ps = 0 v or dv dd . external mclk in 3.25 mw max buf bit = 0. all gains 1 mhz clock 5 mw max buf bit = 1. all gains 1 mhz clock 6.5 mw max buf bit = 0. gain = 32 or 128 @ f clk in = 2.4576 mhz 9.5 mw max buf bit = 1. gain = 32 or 128 @ f clk in = 2.4576 mhz standby (power-down) current 20 20 m a max external mclk in = 0 v or dv dd . typically 10 m a. v dd = +5 v standby (power-down) current 20 10 m a max external mclk in = 0 v or dv dd . typically 5 m a. v dd = +3.3 v notes 1 temperature range as follows: a version, C40 c to +85 c. 2 a calibration is effectively a conversion so these errors will be of the order of the conversion noise shown in tables v to xii . this applies after calibration at the temperature of interest. 3 recalibration at any temperature will remove these drift errors. 4 positive full-scale error includes zero-scale errors (unipolar offset error or bipolar zero error) and applies to both unipolar and bipolar input ranges. 5 full-scale drift includes zero-scale drift (unipolar offset drift or bipolar zero drift) and applies to both unipolar and bipol ar input ranges. 6 gain error does not include zero-scale errors. it is calculated as full-scale errorCunipolar offset error for unipolar ranges a nd full-scale errorCbipolar zero error for bipolar ranges. 7 gain error drift does not include unipolar offset drift/bipolar zero drift. it is effectively the drift of the part if zero sca le calibrations only were performed. 8 these numbers are guaranteed by design and/or characterization. 9 this common-mode voltage range is allowed provided that the input voltage on ain(+) or ain(C) does not go more positive than a v dd + 30 mv or go more nega- tive than agnd C 30 mv. 10 the analog input voltage range on ain(+) is given here with respect to the voltage on ain(C). the absolute voltage on the anal og inputs should not go more posi- tive than av dd + 30 mv or go more negative than agnd C 30 mv. 11 v ref = ref in(+) C ref in(C). 12 these logic output levels apply to the mclk out only when it is loaded with one cmos load. 13 sample tested at +25 c to ensure compliance. 14 after calibration, if the analog input exceeds positive full scale, the converter will output all 1s. if the analog input is le ss than negative full scale, then the device will output all 0s. 15 these calibration and span limits apply provided the absolute voltage on the analog inputs does not exceed av dd + 30 mv or go more negative than agnd C 30 mv. the offset calibration limit applies to both the unipolar zero point and the bipolar zero point. 16 assumes clk bit of setup register is set to correct status corresponding to the master clock frequency. 17 when using a crystal or ceramic resonator across the mclk pins as the clock source for the device, the dv dd current and power dissipation will vary depending on the crystal or resonator type (see clocking and oscillator circuit section). 18 measured at dc and applies in the selected passband. psrr at 50 hz will exceed 120 db with filter notches of 25 hz or 50 hz. ps rr at 60 hz will exceed 120 db with filter notches of 20 hz or 60 hz. 19 psrr depends on gain. gain of 1: 85 db typ; gain of 2: 90 db typ; gains of 32 and 128: 95 db typ. 20 if the external master clock continues to run in standby mode, the standby current increases to 50 m a typical. when using a crystal or ceramic resonator across the mclk pins as the clock source for the device, the internal oscillator continues to run in standby mode and the power dissipatio n depends on the crystal or resonator type (see standby mode section). specifications subject to change without notice. AD7715Cspecifications a (av dd = +3 v to +5 v, dv dd = +3 v to +5 v, ref in(+) = +1.25 v (AD7715-3) or +2.5 v (AD7715-5); ref in(C) = agnd; mclk in = 1 mhz to 2.4576 mhz unless otherwise noted. all specifications t min to t max unless otherwise noted.) C4C rev. b
AD7715 C5C rev. b timing characteristics 1, 2 limit at t min , t max parameter (a version) units conditions/comments f clkin 3, 4 400 khz min master clock frequency: crystal oscillator or externally supplied 2.5 mhz max for specified performance t clk in lo 0.4 t clk in ns min master clock input low time. t clk in = 1/f clk in t clk in hi 0.4 t clk in ns min master clock input high time t 1 500 t clk in ns nom drdy high time t 2 100 ns min reset pulsewidth read operation t 3 0 ns min drdy to cs setup time t 4 120 ns min cs falling edge to sclk rising edge setup time t 5 5 0 ns min sclk falling edge to data valid delay 80 ns max dv dd = +5 v 100 ns max dv dd = +3.3 v t 6 100 ns min sclk high pulsewidth t 7 100 ns min sclk low pulsewidth t 8 0 ns min cs rising edge to sclk rising edge hold time t 9 6 10 ns min bus relinquish time after sclk rising edge 60 ns max dv dd = +5 v 100 ns max dv dd = +3.3 v t 10 100 ns max sclk falling edge to drdy high 7 write operation t 11 120 ns min cs falling edge to sclk rising edge setup time t 12 30 ns min data valid to sclk rising edge setup time t 13 20 ns min data valid to sclk rising edge hold time t 14 100 ns min sclk high pulsewidth t 15 100 ns min sclk low pulsewidth t 16 0 ns min cs rising edge to sclk rising edge hold time notes 1 sample tested at +25 c to ensure compliance. all input signals are specified with tr = tf = 5 ns (10% to 90% of d v dd ) and timed from a voltage level of 1.6 v. 2 see figures 6 and 7. 3 clkin duty cycle range is 45% to 55%. clkin must be supplied whenever the AD7715 is not in standby mode. if no clock is presen t in this case, the device can draw higher current than specified and possibly become uncalibrated. 4 the AD7715 is production tested with f clkin at 2.4576 mhz (1 mhz for some i dd tests). it is guaranteed by characterization to operate at 400 khz. 5 these numbers are measured with the load circuit of figure 1 and defined as the time required for the output to cross the v ol or v oh limits. 6 these numbers are derived from the measured time taken by the data output to change 0.5 v when loaded with the circuit of figur e 1. the measured number is then extrapolated back to remove effects of charging or discharging the 50 pf capacitor. this means that the times quoted in th e timing characteristics are the true bus relinquish times of the part and as such are independent of external bus loading capacitances. 7 drdy returns high after the first read from the device after an output update. the same data can be read again, if required, while drdy is high although care should be taken that subsequent reads do not occur close to the next output update. specifications subject to change without notice. to output pin +1.6v i sink (800 m a at dv dd = +5v 100 m a at dv dd = +3.3v) 50pf i source (200 m a at dv dd = +5v 100 m a at dv dd = +3.3v) figure 1. load circuit for access time and bus relinquish time (dv dd = +3 v to +5.25 v; av dd = +3 v to +5.25 v; agnd = dgnd = 0 v; f clkin = 2.4576 mhz; input logic 0 = 0 v, logic 1 = dv dd , unless otherwise noted)
rev. b AD7715 C6C ordering guide av dd temperature package model supply range options* AD7715an-5 5 v C40 c to +85 c n-16 AD7715ar-5 5 v C40 c to +85 c r-16 AD7715aru-5 5 v C40 c to +85 c ru-16 AD7715an-3 3 v C40 c to +85 c n-16 AD7715ar-3 3 v C40 c to +85 c r-16 AD7715aru-3 3 v C40 c to +85 c ru-16 AD7715achips-5 5 v C40 c to +85 cdie AD7715achips-3 3 v C40 c to +85 cdie eval-AD7715-5eb 5 v evaluation board eval-AD7715-3eb 3 v evaluation board *n = plastic dip; r = soic ru = tssop. absolute maximum ratings* (t a = +25 c unless otherwise noted) av dd to agnd . . . . . . . . . . . . . . . . . . . . . . . . C0.3 v to +7 v av dd to dgnd . . . . . . . . . . . . . . . . . . . . . . . . C0.3 v to +7 v dv dd to agnd . . . . . . . . . . . . . . . . . . . . . . . . C0.3 v to +7 v dv dd to dgnd . . . . . . . . . . . . . . . . . . . . . . . . C0.3 v to +7 v dgnd to agnd . . . . . . . . . . . . . . . . . . . . . . . C0.3 v to +7 v analog input voltage to agnd . . . . . C0.3 v to av dd + 0.3 v reference input voltage to agnd . . . C0.3 v to av dd + 0.3 v digital input voltage to dgnd . . . . . C0.3 v to dv dd + 0.3 v digital output voltage to dgnd . . . . C0.3 v to dv dd + 0.3 v operating temperature range commercial (a version) . . . . . . . . . . . . . . . C40 c to +85 c storage temperature range . . . . . . . . . . . . . C65 c to +150 c junction temperature . . . . . . . . . . . . . . . . . . . . . . . . . +150 c plastic dip package, power dissipation . . . . . . . . . . . 450 mw q ja thermal impedance . . . . . . . . . . . . . . . . . . . . . 105 c/w lead temperature, (soldering, 10 sec) . . . . . . . . . . +260 c soic package, power dissipation . . . . . . . . . . . . . . . . 450 mw q ja thermal impedance . . . . . . . . . . . . . . . . . . . . . . 75 c/w lead temperature, soldering vapor phase (60 sec) . . . . . . . . . . . . . . . . . . . . . . +215 c infrared (15 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . +220 c tssop package, power dissipation . . . . . . . . . . . . . . 450 mw q ja thermal impedance . . . . . . . . . . . . . . . . . . . . . 128 c/w lead temperature, soldering vapor phase (60 sec) . . . . . . . . . . . . . . . . . . . . . . +215 c infrared (15 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . +220 c power dissipation (any package) to +75 c . . . . . . . . 450 mw esd rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . >4000 v *stresses above those listed under absolute maximum ratings may cause perma- nent damage to the device. this is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. exposure to absolute maximum rating conditions for extended periods may affect device reliability. pin configuration dip, soic and tssop 14 13 12 11 16 15 10 9 8 1 2 3 4 7 6 5 top view (not to scale) AD7715 sclk dout din dv dd dgnd mclk in mclk out cs ref in(+) agnd drdy reset avdd ain(+) ain(C) ref in(C)
AD7715 C7C rev. b pin function description pin no. mnemonic function 1 sclk serial clock. logic input. an external serial clock is applied to this input to access serial data from the AD7715. this serial clock can be a continuous clock with all data transmitted in a continuous train of pulses. alternatively, it can be a noncontinuous clock with the information being transmit- ted to the AD7715 in smaller batches of data. 2 mclk in master clock signal for the device. this can be provided in the form of a crystal/resonator or exter- nal clock. a crystal/resonator can be tied across the mclk in and mclk out pins. alterna- tively, the mclk in pin can be driven with a cmos-compatible clock and mclk out left unconnected. the part is specified with clock input frequencies of both 1 mhz and 2.4576 mhz. 3 mclk out when the master clock for the device is a crystal/resonator, the crystal/resonator is connected be- tween mclk in and mclk out. if an external clock is applied to mclk in, mclk out provides an inverted clock signal. this clock can be used to provide a clock source for external circuitry. 4 cs chip select. active low logic input used to select the AD7715. with this input hardwired low, the AD7715 can operate in its three-wire interface mode with sclk, din and dout used to inter- face to the device. cs can be used to select the device in systems with more than one device on the serial bus or as a frame synchronization signal in communicating with the AD7715. 5 reset logic input. active low input which resets the control logic, interface logic, calibration coefficients, digital filter and analog modulator of the part to power-on status. 6av dd analog positive supply voltage, +3.3 v nominal (AD7715-3) or +5 v nominal (AD7715-5). 7 ain(+) analog input. positive input of the programmable gain differential analog input to the AD7715. 8 ain(C) analog input. negative input of the programmable gain differential analog input to the AD7715. 9 ref in(+) reference input. positive input of the differential reference input to the AD7715. the reference input is differential with the provision that ref in(+) must be greater than ref in(C). ref in(+) can lie anywhere between av dd and agnd. 10 ref in(C) reference input. negative input of the differential reference input to the AD7715. the ref in(C) can lie anywhere between av dd and agnd provided ref in(+) is greater than ref in(C). 11 agnd ground reference point for analog circuitry. for correct operation of the AD7715, no voltage on any of the other pins should go more than 30 mv negative with respect to agnd. 12 drdy logic output. a logic low on this output indicates that a new output word is available from the AD7715 data register. the drdy pin will return high upon completion of a read operation of a full output word. if no data read has taken place between output updates, the drdy line will return high for 500 t clk in cycles prior to the next output update. while drdy is high, a read operation should not be attempted or in progress to avoid reading from the data register as it is being updated. the drdy line will return low again when the update has taken place. drdy is also used to indi- cate when the AD7715 has completed its on-chip calibration sequence. 13 dout serial data output with serial data being read from the output shift register on the part. this output shift register can contain information from the setup register, communications register or data regis- ter depending on the register selection bits of the communications register. 14 din serial data input with serial data being written to the input shift register on the part. data from this input shift register is transferred to the setup register or communications register depending on the register selection bits of the communications register. 15 dv dd digital supply voltage, +3.3 v or +5 v nominal. 16 dgnd ground reference point for digital circuitry.
rev. b AD7715 C8C terminology integral nonlinearity this is the maximum deviation of any code from a straight line passing through the endpoints of the transfer function. the end- points of the transfer function are zero-scale (not to be confused with bipolar zero), a point 0.5 lsb below the first code transition (000 . . . 000 to 000 . . . 001) and full-scale, a point 0.5 lsb above the last code transition (111 . . . 110 to 111 . . . 111). the error is expressed as a percentage of full scale. positive full-scale error positive full-scale error is the deviation of the last code transi- tion (111 . . . 110 to 111 . . . 111) from the ideal ain(+) voltage (ain(C) + v ref /gain C3/2 lsbs). it applies to both unipolar and bipolar analog input ranges. unipolar offset error unipolar offset error is the deviation of the first code transition from the ideal ain(+) voltage (ain(C) + 0.5 lsb) when oper- ating in the unipolar mode. bipolar zero error this is the deviation of the midscale transition (0111 . . . 111 to 1000 . . . 000) from the ideal ain(+) voltage (ain(C) C 0.5 lsb) when operating in the bipolar mode. gain error this is a measure of the span error of the adc. it includes full- scale errors but not zero-scale errors. for unipolar input ranges it is defined as (full scale errorCunipolar offset error) while for bipolar input ranges it is defined as (full-scale errorCbipolar zero error). bipolar negative full-scale error this is the deviation of the first code transition from the ideal ain(+) voltage (ain(C) C v ref /gain + 0.5 lsb), when oper- ating in the bipolar mode. positive full-scale overrange positive full-scale overrange is the amount of overhead available to handle input voltages on ain(+) input greater than ain(C) + v ref /gain (for example, noise peaks or excess voltages due to system gain errors in system calibration routines) without intro- ducing errors due to overloading the analog modulator or over- flowing the digital filter. negative full-scale overrange this is the amount of overhead available to handle voltages on ain(+) below ain(C) Cv ref /gain without overloading the analog modulator or overflowing the digital filter. note that the analog input will accept negative voltage peaks even in the uni- polar mode provided that ain(+) is greater than ain(C) and greater than agnd C 30 mv. offset calibration range in the system calibration modes, the AD7715 calibrates its offset with respect to the analog input. the offset calibration range specification defines the range of voltages that the AD7715 can accept and still calibrate offset accurately. full-scale calibration range this is the range of voltages that the AD7715 can accept in the system calibration mode and still calibrate full scale correctly. input span in system calibration schemes, two voltages applied in sequence to the AD7715s analog input define the analog input range. the input span specification defines the minimum and maxi- mum input voltages from zero to full scale that the AD7715 can accept and still calibrate gain accurately. on-chip registers the part contains four on-chip registers which can be accessed by via the serial port on the part. the first of these is a comm unica- tions register that decides whether the next operation is a read or write operation and also decides which register the read or write operation accesses. all communications to the part must start with a write operation to the communications register. after powe r- on or reset, the device expects a write to its communications register. the data written to this register determines whether th e next operation to the part is a write or a read operation and also determines to which register this read or write operation oc curs. therefore, write access to any of the other registers on the part starts with a write operation to the communications register fol- lowed by a write to the selected register. a read operation from any register on the part (including the communications registe r itself and the output data register) starts with a write operation to the communications register followed by a read operation from th e selected register. the communication register also controls the standby mode and the operating gain of the part. the drdy status is also available by reading from the communications register. the second register is a setup register that determines calibrat ion modes, filter selection and bipolar/unipolar operation. the third register is the data register from which the output data from the part is accessed. the final register is a test register that is accessed when testing the device. it is advised that the user d oes not attempt to access or change the contents of the test register as it may lead to unspecified operation of the device. the regist ers are discussed in more detail in the following sections.
AD7715 C9C rev. b communications register (rs1, rs0 = 0, 0) the communications register is an eight-bit register from which data can either be read or to which data can be written. all co m- munications to the part must start with a write operation to the communications register. the data written to the communication s register determines whether the next operation is a read or write operation and to which register this operation takes place. o nce the subsequent read or write operation to the selected register is complete, the interface returns to where it expects a write oper ation to the communications register. this is the default state of the interface, and on power-up or after a reset , the AD7715 is in this default state waiting for a write operation to the communications register. in situations where the interface sequence is lost, if a write operation to the device of sufficient duration (containing at least 32 serial clock cycles) takes place with din high, th e AD7715 returns to this default state. table i outlines the bit designations for the communications register. table i. communications register 0/ drdy zero rs1 rs0 r/ w stby g1 g0 0/ drdy for a write operation, a 0 must be written to this bit so that the write operation to the communications reg- ister actually takes place. if a 1 is written to this bit, the part will not clock on to subsequent bits in the regis- ter. it will stay at this bit location until a 0 is written to this bit. once a 0 is written to this bit, the next 7 bits will be loaded to the communications register. for a read operation, this bit provides the status of the drdy flag from the part. the status of this bit is the same as the drdy output pin. zero for a write operation, a 0 must be written to this bit for correct operation of the part. failure to do this will result in unspecified operation of the device. for a read operation, a 0 will be read back from this bit location. rs1C rs0 register selection bits. these bits select to which one of four on-chip registers the next read or write opera- tion takes place as shown in table ii along with the register size. when the read or write to the selected regis- ter is complete, the part returns to where it is waiting for a write operation to the communications register. it does not remain in a state where it will continue to access the selected register. r/ w read/write select. this bit selects whether the next operation is a read or write operation to the selected register. a 0 indicates a write cycle as the next operation to the appropriate register, while a 1 indicates a read operation from the appropriate register. table ii. register selection rs1 rs0 register register size 0 0 communications register 8 bits 0 1 setup register 8 bits 1 0 test register 8 bits 1 1 data register 16 bits stby standby. writing a 1 to this bit puts the part in its standby or power-down mode. in this mode, the part consumes only 10 m a of power supply current. the part retains its calibration and control word information when in standby. writing a 0 to this bit places the part in its normal operating mode. the default value for this bit after power-on or reset is 0. g2 g1 gain setting 00 1 01 2 10 32 1 1 128
rev. b AD7715 C10C setup register (rs1, rs0 = 0, 1); power on/reset status: 28 hex the setup register is an eight-bit register from which data can either be read or to which data can be written. this register c ontrols the setup which the device is to operate in such as the calibration mode, output rate, unipolar/bipolar operation etc. table ii i out- lines the bit designations for the setup register. table iii. setup register md1 md0 clk fs1 fs0 b /u buf fsync md1 md0 operating mode 0 0 normal mode; this is the normal mode of operation of the device whereby the device is performing normal conversions. this is the default condition of these bits after power-on or reset. 0 1 self-calibration; this activates self-calibration on the part. this is a one step calibration sequence and when complete the part returns to normal mode with md1 and md0 returning to 0, 0. the drdy output or bit goes high when calibration is initiated and returns low when this self-calibration is complete and a new valid word is available in the data register. the zero-scale calibration is performed at the selected gain on internally shorted (zeroed) inputs and the full-scale calibration is performed at the selected gain on an internally generated v ref /selected gain. 1 0 zero-scale system calibration; this activates zero-scale system calibration on the part. calibration is per- formed at the selected gain on the input voltage provided at the analog input during this calibration sequence. this input voltage should remain stable for the duration of the calibration. the drdy output or bit goes high when calibration is initiated and returns low when this zero-scale calibration is complete and a new valid word is available in the data register. at the end of the calibration, the part returns to normal mode with md1 and md0 returning to 0, 0. 1 1 full-scale system calibration; this activates full-scale system calibration on the part. calibration is per- formed at the selected gain on the input voltage provided at the analog input during this calibration sequence. this input voltage should remain stable for the duration of the calibration. once again, the drdy output or bit goes high when calibration is initiated and returns low when this full-scale calibration is complete and a new valid word is available in the data register. at the end of the calibration, the part returns to normal mode with md1 and md0 returning to 0, 0. clk clock bit. this bit should be set in accordance with the operating frequency of the AD7715. if the device has a master clock frequency of 2.4576 mhz, then this bit should be set to a 1. if the device has a master clock frequency of 1 mhz, then this bit should be set to a 0. this bit sets up the correct scaling currents for a given master clock and also chooses (along with fs1 and fs0) the output update rate for the device. if this bit is not set correctly for the master clock frequency of the device, then the device may not operate to specifica- tion. the default value for this bit after power-on or reset is 1. fs1, fs0 filter selection bits. along with the clk bit, fs1 and fs0 determine the output update rate, filter first notch and C3 db frequency as outlined in table iv. the on-chip digital filter provides a sinc 3 (or (sinx/x) 3 ) filter response. in association with the gain selection, it also determines the output noise (and hence the resolution) of the device. changing the filter notch frequency, as well as the selected gain, impacts resolution. tables v through xii show the effect of the filter notch frequency and gain on the output noise and effective resolution of the part. the output data rate (or effective conversion time) for the device is equal to the fre- quency selected for the first notch of the filter. for example, if the first notch of the filter is selected at 50 hz then a new word is available at a 50 hz rate or every 20 ms. if the first notch is at 500 hz, a new word is available every 2 ms. the default value for these bits is 1, 0. the settling-time of the filter to a full-scale step input change is worst case 4 1/(output data rate). for example, with the first filter notch at 50 hz, the settling time of the filter to a full-scale step input change is 80 ms max. if the first notch is at 500 hz, the settling time of the filter to a full-scale input step is 8 ms max. this settling-time can be reduced to 3 1/(output data rate) by synchronizing the step input change to a reset of the digital filter. in other words, if the step input takes place with the fsync bit high, the settling- time time will be 3 1/(output data rate) from when fsync returns low. the C3 db frequency is determined by the programmed first notch frequency according to the relationship: filter C3 db frequency = 0.262 filter first notch frequency.
AD7715 C11C rev. b table iv. output update rates clk* fs1 fs0 output update rate C3 db filter cutoff 0 0 0 20 hz 5.24 hz 0 0 1 25 hz 6.55 hz 0 1 0 100 hz 26.2 hz 0 1 1 200 hz 52.4 hz 1 0 0 50 hz 13.1 hz 1 0 1 60 hz 15.7 hz default status 1 1 0 250 hz 65.5 hz 1 1 1 500 hz 131 hz *assumes correct clock frequency at mclk in pin b /u bipolar/unipolar operation. a 0 in this bit selects bipolar operation. this is the default (power-on or reset ) status of this bit. a 1 in this bit selects unipolar operation. buf buffer control. with this bit low, the on-chip buffer on the analog input is shorted out. with the buffer shorted out, the current flowing in the av dd line is reduced to 250 m a (all gains at f clk in = 1 mhz and gain of 1 or 2 at f clk in = 2.4576 mhz) or 500 m a (gains of 32 and 128 @ f clk in = 2.4576 mhz) and the output noise from the part is at its lowest. when this bit is high, the on-chip buffer is in series with the analog input allowing the input to handle higher source impedances. fsync filter synchronization. when this bit is high, the nodes of the digital filter, the filter control logic and the calibration control logic are held in a reset state and the analog modulator is also held in its reset state. when this bit goes low, the modulator and filter start to process data and a valid word is available in 3 1/(output update rate), i.e., the settling-time of the filter. this fsync bit does not affect the digital interface and does not reset the drdy output if it is low. test register (rs1, rs0 = 1, 0) the part contains a test register which is used in testing the device. the user is advised not to change the status of any of t he bits in this register from the default (power-on or reset) status of all 0s as the part will be placed in one of its test modes and will not operate correctly. if the part enters one of its test modes, exercising reset will exit the part from the mode. an alterna- tive scheme for getting the part out of one of its test modes, is to reset the interface by writing 32 successive 1s to the par t and then load all 0s to the test register. data register (rs1, rs0 = 1, 1) the data register on the part is a read-only 16-bit register which contains the most up-to-date conversion result from the AD7715. if the communications register data sets up the part for a write operation to this register, a write operation must act u- ally take place to return the part to where it is expecting a write operation to the communications register (the default stat e of the interface). however, the 16 bits of data written to the part will be ignored by the AD7715. output noise AD7715-5 table v shows the AD7715-5 output rms noise for the selectable notch and C3 db frequencies for the part, as selected by fs1 and fs0 of the setup register. the numbers given are for the bipolar input ranges with a v ref of +2.5 v. these numbers are typical and are generated at a differential analog input voltage of 0 v with the part used in unbuffered mode (buf bit of the setup register = 0). table vi meanwhile shows the output peak-to-peak noise for the selectable notch and C3 db frequencies for the part. it is important to note that these numbers represent the resolution for which there will be no code flicker. they are not calcu lated based on rms noise but on peak-to-peak noise . the numbers given are for the bipolar input ranges with a v ref of +2.5 v and for the buf bit of the setup register = 0. these numbers are typical, are generated at an analog input voltage of 0 v and are rounded to the nearest lsb. meanwhile, table vii and table viii show rms noise and peak-to-peak resolution respectively with the AD7715-5 operating under the same conditions as above except that now the part is operating in buffered mode (buf bit of the setup register = 1).
rev. b AD7715 C12C table v. output rms noise vs. gain and output update rate for AD7715-5 (unbuffered mode) filter first notch & o/p data rate C3 db frequency typical output rms noise in m v mclk in = mclk in = mclk in = mclk in = 2.4576 mhz 1 mhz 2.4576 mhz 1 mhz gain = 1 gain = 2 gain = 32 gain = 128 50 hz 20 hz 13.1 hz 5.24 hz 3.8 1.9 0.6 0.52 60 hz 25 hz 15.72 hz 6.55 hz 4.8 2.4 0.6 0.62 250 hz 100 hz 65.5 hz 26.2 hz 103 45 3.0 1.6 500 hz 200 hz 131 hz 52.4 hz 530 250 18 5.5 table vi. peak-to-peak resolution vs. gain and output update rate for AD7715-5 (unbuffered mode) filter first notch & o/p data rate C3 db frequency typical peak-to-peak resolution in bits mclk in = mclk in = mclk in = mclk in = 2.4576 mhz 1 mhz 2.4576 mhz 1 mhz gain = 1 gain = 2 gain = 32 gain = 128 50 hz 20 hz 13.1 hz 5.24 hz 16 16 16 14 60 hz 25 hz 15.72 hz 6.55 hz 16 16 16 13 250 hz 100 hz 65.5 hz 26.2 hz 13 13 13 12 500 hz 200 hz 131 hz 52.4 hz 10 10 10 10 table vii. output rms noise vs. gain and output update rate for AD7715-5 (buffered mode) filter first notch & o/p data rate C3 db frequency typical output rms noise in m v mclk in = mclk in = mclk in = mclk in = 2.4576 mhz 1 mhz 2.4576 mhz 1 mhz gain = 1 gain = 2 gain = 32 gain = 128 50 hz 20 hz 13.1 hz 5.24 hz 4.3 2.2 0.9 0.9 60 hz 25 hz 15.72 hz 6.55 hz 5.1 3.1 1.0 1.0 250 hz 100 hz 65.5 hz 26.2 hz 103 50 3.9 2.1 500 hz 200 hz 131 hz 52.4 hz 550 280 18 6 table viii. peak-to-peak resolution vs. gain and output update rate for AD7715-5 (buffered mode) filter first notch & o/p data rate C3 db frequency typical peak-to-peak resolution in bits mclk in = mclk in = mclk in = mclk in = 2.4576 mhz 1 mhz 2.4576 mhz 1 mhz gain = 1 gain = 2 gain = 32 gain = 128 50 hz 20 hz 13.1 hz 5.24 hz 16 16 15 13 60 hz 25 hz 15.72 hz 6.55 hz 16 16 15 13 250 hz 100 hz 65.5 hz 26.2 hz 13 13 13 12 500 hz 200 hz 131 hz 52.4 hz 10 10 10 10 AD7715-3 table ix shows the AD7715-3 output rms noise for the selectable notch and C3 db frequencies for the part, as selected by fs1 an d fs0 of the setup register. the numbers given are for the bipolar input ranges with a v ref of +1.25 v. these numbers are typical and are generated at an analog input voltage of 0 v with the part used in unbuffered mode (buf bit of the setup register = 0). table x meanwhile shows the output peak-to-peak noise for the selectable notch and C3 db frequencies for the part. it is important to note that these numbers represent the resolution for which there will be no code flicker. they are not calculated based on rms noise but on peak- to-peak noise . the numbers given are for the bipolar input ranges with a v ref of +1.25 v and for the buf bit of the setup register = 0. these numbers are typical, are generated at an analog input voltage of 0 v and are rounded to the nearest lsb. meanwhile, table xi and table xii show rms noise and peak-to-peak resolution respectively with the AD7715-3 operating under the same conditions as above except that now the part is operating in buffered mode (buf bit of the setup register = 1).
AD7715 C13C rev. b table ix. output rms noise vs. gain and output update rate for AD7715-3 (unbuffered mode) filter first notch & o/p data rate C3 db frequency typical output rms noise in m v mclk in = mclk in = mclk in = mclk in = 2.4576 mhz 1 mhz 2.4576 mhz 1 mhz gain = 1 gain = 2 gain = 32 gain = 128 50 hz 20 hz 13.1 hz 5.24 hz 3.0 1.7 0.7 0.65 60 hz 25 hz 15.72 hz 6.55 hz 3.4 2.1 0.7 0.7 250 hz 100 hz 65.5 hz 26.2 hz 45 20 2.2 1.6 500 hz 200 hz 131 hz 52.4 hz 270 135 9.7 3.3 table x. peak-to-peak resolution vs. gain and output update rate for AD7715-3 (unbuffered mode) filter first notch & o/p data rate C3 db frequency typical peak-to-peak resolution in bits mclk in = mclk in = mclk in = mclk in = 2.4576 mhz 1 mhz 2.4576 mhz 1 mhz gain = 1 gain = 2 gain = 32 gain = 128 50 hz 20 hz 13.1 hz 5.24 hz 16 16 14 12 60 hz 25 hz 15.72 hz 6.55 hz 16 16 14 12 250 hz 100 hz 65.5 hz 26.2 hz 13 13 13 11 500 hz 200 hz 131 hz 52.4 hz 11 11 10 10 table xi. output rms noise vs. gain and output update rate for AD7715-3 (buffered mode) filter first notch & o/p data rate C3 db frequency typical output rms noise in m v mclk in = mclk in = mclk in = mclk in = 2.4576 mhz 1 mhz 2.4576 mhz 1 mhz gain = 1 gain = 2 gain = 32 gain = 128 50 hz 20 hz 13.1 hz 5.24 hz 4.5 2.4 0.9 0.9 60 hz 25 hz 15.72 hz 6.55 hz 5.1 2.9 0.9 1.0 250 hz 100 hz 65.5 hz 26.2 hz 50 25 2.6 2 500 hz 200 hz 131 hz 52.4 hz 270 135 9.7 3.3 table xii. peak-to-peak resolution vs. gain and output update rate for AD7715-3 (buffered mode) filter first notch & o/p data rate C3 db frequency typical peak-to-peak resolution in bits mclk in = mclk in = mclk in = mclk in = 2.4576 mhz 1 mhz 2.4576 mhz 1 mhz gain = 1 gain = 2 gain = 32 gain = 128 50 hz 20 hz 13.1 hz 5.24 hz 16 16 14 12 60 hz 25 hz 15.72 hz 6.55 hz 16 16 14 12 250 hz 100 hz 65.5 hz 26.2 hz 13 13 12 11 500 hz 200 hz 131 hz 52.4 hz 10 11 10 10
rev. b AD7715 C14C calibration sequences the AD7715 contains a number of calibration options as outlined previously. table xiii summarizes the calibration types, the op - erations involved and the duration of the operations. there are two methods of determining the end of calibration. the first is to monitor when drdy returns low at the end of the sequence. drdy not only indicates when the sequence is complete but also that the part has a valid new sample in its data register. this valid new sample is the result of a normal conversion which follows the cali- bration sequence. the second method of determining when calibration is complete is to monitor the md1 and md0 bits of the setup register. when these bits return to 0, 0 following a calibration command, it indicates that the calibration sequence is c om- plete. this method does not give any indication of there being a valid new result in the data register. however, it gives an ea rlier indication than drdy that calibration is complete. the duration to when the mode bits (md1 and md0) return to 0, 0 represents the duration of the calibration carried out. the sequence to when drdy goes low also includes a normal conversion and a pipeline delay, t p , to correctly scale the results of this first conversion. t p will never exceed 2000 t clk in . the time for both methods is given in the table. table xiii. calibration sequences calibration type md1, md0 calibration sequence duration to mode bits duration to drdy self calibration 0, 1 internal zs cal @ selected gain + 6 1/output rate 9 1/output rate + t p internal fs cal @ selected gain zs system calibration 1, 0 zs cal on ain @ selected gain 3 1/output rate 4 1/output rate + t p fs system calibration 1, 1 fs cal on ain @ selected gain 3 1/output rate 4 1/output rate + t p circuit description the AD7715 is a sigma-delta a/d converter with on-chip digital filtering, intended for the measurement of wide dynamic range, low frequency signals such as those in industrial control or pro- cess control applications. it contains a sigma-delta (or charge- balancing) adc, a calibration microcontroller with on-chip static ram, a clock oscillator, a digital filter and a bidirectional serial communications port. the part consumes only 450 m a of power supply current, making it ideal for battery-powered or loop-powered instruments. the part comes in two versions, the AD7715-5 which is specified for operation from a nominal +5 v analog supply (av dd ) and the AD7715-3 which is speci- fied for operation from a nominal +3.3 v analog supply. both versions can be operated with a digital supply (dv dd ) voltage of +3.3 v or +5 v. the part contains a programmable-gain fully differential analog input channel. the selectable gains on this input are 1, 2, 32 and 128 allowing the part to accept unipolar signals of between 0 mv to +20 mv and 0 v to +2.5 v or bipolar signals in the range from 20 mv to 2.5 v when the reference input voltage equals +2.5 v. with a reference voltage of +1.25 v, the input ranges are from 0 mv to +10 mv to 0 v to +1.25 v in unipolar mode and from 10 mv to 1.25 v in bipolar mode. note that the bipolar ranges are with respect to ain(C) and not with re- spect to agnd. the input signal to the analog input is continuously sampled at a rate determined by the frequency of the master clock, mclk in, and the selected gain. a charge-balancing a/d converter (sigma-delta modulator) converts the sampled signal into a digital pulse train whose duty cycle contains the digital information. the programmable gain function on the analog input is also incorporated in this sigma-delta modulator with the input sampling frequency being modified to give the higher gains. a sinc 3 digital low-pass filter processes the output of the sigma-delta modulator and updates the output register at a rate determined by the first notch frequency of this filter. the out- put data can be read from the serial port randomly or periodi- cally at any rate up to the output register update rate. the first notch of this digital filter (and hence its C3 db frequency) can be programmed via the setup register bits fs0 and fs1. with a master clock frequency of 2.4576 mhz, the programmable range for this first notch frequency is from 50 hz to 500 hz giving a programmable range for the C3 db frequency of 13.1 hz to 131 hz. with a master clock frequency of 1 mhz, the programmable range for this first notch frequency is from 20 hz to 200 hz giving a programmable range for the C3 db frequency of 5.24 hz to 52.4 hz. the basic connection diagram for the AD7715-5 is shown in figure 2. this shows both the av dd and dv dd pins of the AD7715 being driven from the analog +5 v supply. some applications will have av dd and dv dd driven from separate supplies. an ad780, precision +2.5 v reference, provides the reference source for the part. on the digital side, the part is configured for three-wire operation with cs tied to dgnd. a quartz crystal or ceramic resonator provides the master clock source for the part. in most cases, it will be necessary to connect capacitors on the crystal or resonator to ensure that it does not oscillate at overtones of its fundamental operating fre- quency. the values of capacitors will vary depending on the manufacturers specifications.
AD7715 C15C rev. b c samp must be charged through r sw and through any external source impedances every input sample cycle. therefore, in unbuffered mode, source impedances mean a longer charge time for c samp , and this may result in gain errors on the part. table xiv shows the allowable external resistance/capacitance values, for unbuffered mode, such that no gain error to the 16-bit level is introduced on the part. note that these capacitances are total capacitances on the analog input, external capacitance plus 10 pf capacitance from the pins and lead frame of the device. table xiv. external r, c combination for no 16-bit gain error (unbuffered mode only) gain external capacitance (pf) 10 50 100 500 1000 5000 1 152 k w 53.9 k w 31.4 k w 8.4 k w 4.76 k w 1.36 k w 2 75.1 k w 26.6 k w 15.4 k w 4.14 k w 2.36 k w 670 w 32 16.7 k w 5.95 k w 3.46 k w 924 w 526 w 150 w 128 16.7 k w 5.95 k w 3.46 k w 924 w 526 w 150 w in buffered mode, the analog inputs look into the high imped- ance inputs stage of the on-chip buffer amplifier. c samp is charged via this buffer amplifier such that source impedances do not affect the charging of c samp . this buffer amplifier has an offset leakage current of 1 na. in this buffered mode, large source impedances result in a small dc offset voltage developed across the source impedance but not in a gain error. input sample rate the modulator sample frequency for the AD7715 remains at f clk in /128 (19.2 khz @ f clk in = 2.4576 mhz) regardless of the selected gain. however, gains greater than 1 are achieved by a combination of multiple input samples per modulator cycle and a scaling of the ratio of reference capacitor to input capaci- tor. as a result of the multiple sampling, the input sample rate of the device varies with the selected gain (see table xv). in buffered mode, the input is buffered before the input sampling table xv. input sampling frequency vs. gain gain input sampling freq (f s ) 1f clk in /64 (38.4 khz @ f clk in = 2.4576 mhz) 22 f clk in /64 (76.8 khz @ f clk in = 2.4576 mhz) 32 8 f clk in /64 (307.2 khz @ f clk in = 2.4576 mhz) 128 8 f clk in /64 (307.2 khz @ f clk in = 2.4576 mhz) capacitor. in unbuffered mode, where the analog input looks directly into the sampling capacitor, the effective input imped- ance is 1/c samp f s where c samp is the input sampling capaci- tance and f s is the input sample rate. bipolar/unipolar inputs the analog input on the AD7715 can accept either unipolar or bipolar input voltage ranges. bipolar input ranges do not imply that the part can handle negative voltages on its analog input since the analog input cannot go more negative than C30 mv to ensure correct operation of the part. the input channel is fully differential. as a result, the voltage to which the unipolar and bipolar signals on the ain(+) input are referenced is the voltage on the respective ain(C) input. for example, if ain(C) is +2.5 v and the AD7715 is configured for unipolar operation sclk mclk in dgnd dv dd mclk out din dout agnd ain(+) ain(C) ref in(+) ref in(C) av dd AD7715 0.1 m f analog ground differential analog input digital ground 0.1 m f 10 m f v out v in gnd ad780 analog +5v supply data ready receive (read) serial data serial clock crystal or ceramic resonator +5v 0.1 m f 10 m f analog +5v supply reset cs drdy figure 2. AD7715-5 basic connection diagram analog input analog input ranges the AD7715 contains a differential analog input pair ain(+) and ain(C). this input pair provides a programmable-gain, differential input channel which can handle either unipolar or bipolar input signals. it should be noted that the bipolar input signals are referenced to the respective ain(C) input of the input pair. in unbuffered mode, the common-mode range of the input is from agnd to av dd provided that the absolute value of the analog input voltage lies between agnd C 30 mv and av dd + 30 mv. this means that in unbuffered mode the part can handle both unipolar and bipolar input ranges for all gains. in buffered mode, the analog inputs can handle much larger source impedances but the absolute input voltage range is re- stricted to between agnd + 50 mv to av dd C 1.5 v which also places restrictions on the common-mode range. this means that in buffered mode there are some restrictions on the allow- able gains for bipolar input ranges. care must be taken in set- ting up the common-mode voltage and input voltage range so that the above limits are not exceeded, otherwise there will be a degradation in linearity performance. in unbuffered mode, the analog inputs look directly into the input sampling capacitor, c samp . the dc input leakage current in this unbuffered mode is 1 na maximum. as a result, the analog inputs see a dynamic load that is switched at the input sample rate (see figure 3). this sample rate depends on master clock frequency and selected gain. c samp is charged to ain(+) and discharged to ain(C) every input sample cycle. the effec- tive on-resistance of the switch, r sw , is typically 7 k w . high impedance 1g v r sw (7k v typ) c samp (10pf ) v bias switching frequency depends on f clkin and selected gain ain(+) ain(C) figure 3. unbuffered analog input structure
rev. b AD7715 C16C digital filtering the AD7715 contains an on-chip low-pass digital filter that processes the output of the parts sigma-delta modulator. there- fore, the part not only provides the analog-to-digital conversion function but it also provides a level of filtering. there are a number of system differences when the filtering function is provided in the digital domain rather than the analog domain and the user should be aware of these. first, since digital filtering occurs after the a-to-d conversion process, it can remove noise injected during the conversion process. analog filtering cannot do this. also, the digital filter can be made programmable far more readily than an analog filter. depending on the digital filter design, this gives the user the capability of programming cutoff frequency and output update rate. on the other hand, analog filtering can remove noise superim- posed on the analog signal before it reaches the adc. digital filtering cannot do this and noise peaks riding on signals near full scale have the potential to saturate the analog modulator and digital filter, even though the average value of the signal is within limits. to alleviate this problem, the AD7715 has over- range headroom built into the sigma-delta modulator and digital filter which allows overrange excursions of 5% above the analog input range. if noise signals are larger than this, consideration should be given to analog input filtering, or to reducing the input channel voltage so that its full scale is half that of the analog input channel full scale. this will provide an overrange capability greater than 100% at the expense of reducing the dynamic range by 1 bit (50%). in addition, the digital filter does not provide any rejection at integer multiples of the digital filters sample frequency. how- ever, the input sampling on the part provides attenuation at multiples of the digital filters sampling frequency so that the unattenu-ated bands actually occur around multiples of the sampling frequency f s (as defined in table xv). thus the unat- tenuated bands occur at n f s (where n = 1, 2, 3. . . ). at these frequencies, there are frequency bands, f 3 db wide (f 3 db is the cutoff frequency of the digital filter) at either side where noise passes unattenuated to the output. filter characteristics the AD7715s digital filter is a low-pass filter with a (sinx/x) 3 response (also called sinc 3 ). the transfer function for this filter is described in the z-domain by: and in the frequency domain by: where n is the ratio of the modulator rate to the output rate and f mod is the modulator rate. with a gain of 2 and a v ref of +2.5 v, the input voltage range on the ain(+) input is +2.5 v to +3.75 v. if ain(C) is +2.5 v and the AD7715 is configured for bipolar mode with a gain of 2 and a v ref of +2.5 v, the analog input range on the ain(+) input is +1.25 v to +3.75 v (i.e., 2.5 v 1.25 v). if ain(C) is at agnd, the part cannot be configured for bipolar ranges in excess of 30 mv. bipolar or unipolar options are chosen by programming the b /u bit of the setup register. this programs the channel for either unipolar or bipolar operation. programming the channel for either unipolar or bipolar operation does not change any of the input signal conditioning; it simply changes the data output coding and the points on the transfer function where calibra- tions occur. reference input the AD7715s reference inputs, ref in(+) and ref in(C), provide a differential reference input capability. the common- mode range for these differential inputs is from agnd to av dd . the nominal reference voltage, v ref (ref in(+) C ref in(C)), for specified operation is +2.5 v for the AD7715-5 and +1.25 v for the AD7715-3. the part is functional with v ref voltages down to 1 v but with degraded performance as the output noise will, in terms of lsb size, be larger. ref in(+) must always be greater than ref in(C) for correct operation of the AD7715. both reference inputs provide a high impedance, dynamic load similar to the analog inputs in unbuffered mode. the maximum dc input leakage current is 1 na over temperature and source resistance may result in gain errors on the part. in this case, the sampling switch resistance is 5 k w typ and the reference capaci- tor (c ref ) varies with gain. the sample rate on the reference inputs is f clk in /64 and does not vary with gain. for gains of 1 and 2, c ref is 8 pf; for a gain of 32, it is 4.25 pf, and for a gain of 128, it is 3.3125 pf. the output noise performance outlined in tables v through xii is for an analog input of 0 v which effectively removes the effect of noise on the reference. to obtain the same noise performance as shown in the noise tables over the full input range requires a low noise reference source for the AD7715. if the reference noise in the bandwidth of interest is excessive, it will degrade the performance of the AD7715. in applications where the excitation voltage for the bridge transducer on the analog input also derives the reference voltage for the part, the effect of the noise in the excitation voltage will be removed as the application is ratiometric. recommended reference voltage sources for the AD7715-5 include the ad780, ref43 and ref192, while the recommended reference sources for the AD7715-3 include the ad589 and ad1580. it is generally recommended to decouple the output of these references in order to further reduce the noise level. hz n z z n () = ? ? - 11 1 1 3 |()| hf n s in n f f sin f f s s = ? ? ? ? ? ? ? ? 1 3 p p
AD7715 C17C rev. b figure 4 shows the filter frequency response for a cutoff fre- quency of 15.72 hz which corresponds to a first filter notch frequency of 60 hz. the plot is shown from dc to 390 hz. this response is repeated at either side of the digital filters sample frequency and at either side of multiples of the filters sample frequency. frequency C hz 0 C40 C60 C80 C100 C120 C140 C160 C180 C200 C220 C20 C240 360 0 300 180 120 60 240 gain C db figure 4. frequency response of AD7715 filter the response of the filter is similar to that of an averaging filter but with a sharper roll-off. the output rate for the digital filter corresponds with the positioning of the first notch of the filters frequency response. thus, for the plot of figure 4 where the output rate is 60 hz, the first notch of the filter is at 60 hz. the notches of this (sinx/x) 3 filter are repeated at multiples of the first notch. the filter provides attenuation of better than 100 db at these notches. the cutoff frequency of the digital filter is determined by the value loaded to bits fs0 to fs1 in the setup register. pro- gramming a different cutoff frequency via fs0 and fs1 does not alter the profile of the filter response; it changes the frequency of the notches. the output update of the part and the frequency of the first notch correspond. since the AD7715 contains this on-chip, low-pass filtering, there is a settling time associated with step function inputs and data on the output will be invalid after a step change until the settling time has elapsed. the settling time depends upon the output rate chosen for the filter. the settling time of the filter to a full-scale step input can be up 4 times the output data period. for a synchronized step input (using the fsync func- tion), the settling time is 3 times the output data period. post-filtering the on-chip modulator provides samples at a 19.2 khz output rate with f clk in at 2.4576 mhz. the on-chip digital filter decimates these samples to provide data at an output rate which corresponds to the programmed output rate of the filter. since the output data rate is higher than the nyquist criterion, the output rate for a given bandwidth will satisfy most application requirements. however, there may be some applications which require a higher data rate for a given bandwidth and noise per- formance. applications that need this higher data rate will require some post-filtering following the digital filter of the AD7715. for example, if the required bandwidth is 7.86 hz but the re- quired update rate is 100 hz, the data can be taken from the AD7715 at the 100 hz rate giving a C3 db bandwidth of 26.2 hz. post-filtering can be applied to this to reduce the bandwidth and output noise, to the 7.86 hz bandwidth level, while maintaining an output rate of 100 hz. post-filtering can also be used to reduce the output noise from the device for bandwidths below 13.1 hz. at a gain of 128 and a bandwidth of 13.1 hz, the output rms noise is 520 nv. this is essentially device noise or white noise and since the input is chopped, the noise has a primarily flat frequency response. by reducing the bandwidth below 13.1 hz, the noise in the result- ant passband can be reduced. a reduction in bandwidth by a factor of 2 results in a reduction of approximately 1.25 in the output rms noise. this additional filtering will result in a longer settling time. analog filtering the digital filter does not provide any rejection at integer mul- tiples of the modulator sample frequency, as outlined earlier. however, due to the AD7715s high oversampling ratio, these bands occupy only a small fraction of the spectrum and most broadband noise is filtered. this means that the analog filtering requirements in front of the AD7715 are considerably reduced versus a conventional converter with no on-chip filtering. in addition, because the parts common-mode rejection perfor- mance of 95 db extends out to several khz, common-mode noise in this frequency range will be substantially reduced. depending on the application, however, it may be necessary to provide attenuation prior to the AD7715 in order to eliminate unwanted frequencies from these bands which the digital filter will pass. it may also be necessary in some applications to pro- vide analog filtering in front of the AD7715 to ensure that dif- ferential noise signals outside the band of interest do not saturate the analog modulator. if passive components are placed in front of the AD7715, in unbuffered mode, care must be taken to ensure that the source impedance is low enough so as not to introduce gain errors in the system. this significantly limits the amount of passive anti- aliasing filtering which can be provided in front of the AD7715 when it is used in unbuffered mode. however, when the part is used in buffered mode, large source impedances will simply result in a small dc offset error (a 10 k w source resistance will cause an offset error of less than 10 m v). therefore, if the sys- tem requires any significant source impedances to provide pas- sive analog filtering in front of the AD7715, it is recommended that the part be operated in buffered mode. calibration the AD7715 provides a number of calibration options that can be programmed via the md1 and md0 bits of the setup regis- ter. the different calibration options are outlined in the setup register and calibration sequences sections. a calibration cycle may be initiated at any time by writing to these bits of the setup register. calibration on the AD7715 removes offset and gain errors from the device. a calibration routine should be initiated on the device whenever there is a change in the ambient operat- ing temperature or supply voltage. it should also be initiated if there is a change in the selected gain, filter notch or bipolar/ unipolar input range. the AD7715 offers self-calibration and system-calibration facili- ties. for full calibration to occur on the selected channel, the on-chip microcontroller must record the modulator output for two different input conditions. these are zero-scale and
rev. b AD7715 C18C full-scale points. these points are derived by performing a conversion on the different input voltages provided to the input of the modulator during calibration. as a result, the accuracy of the calibration can only be as good as the noise level that it provides in normal mode. the result of the zero-scale calibra- tion conversion is stored in the zero-scale calibration register while the result of the full-scale calibration conversion is stored in the full-scale calibration register. with these read- ings, the on-chip microcontroller can calculate the offset and the gain slope for the input to output transfer function of the con- verter. internally, the part works with a resolution of 33 bits to determine its conversion result of 16 bits. self-calibration a self-calibration is initiated on the AD7715 by writing the appropriate values (0, 1) to the md1 and md0 bits of the setup register. in the self-calibration mode with a unipolar input range, the zero-scale point used in determining the cali- bration coefficients is with the inputs of the differential pair internally shorted on the part (i.e., ain(+) = ain(C) = internal bias voltage). the pga is set for the selected gain (as per g1 and g0 bits in the communications register) for this zero-scale calibration conversion. the full-scale calibration conversion is performed at the selected gain on an internally generated voltage of v ref /selected gain. the duration time for the calibration is 6 1/output rate. this is made up of 3 1/output rate for the zero-scale calibration and 3 1/output rate for the full-scale calibration. at this time the md1 and md0 bits in the setup register return to 0, 0. this gives the earliest indication that the calibration sequence is complete. the drdy line goes high when calibration is initi- ated and does not return low until there is a valid new word in the data register. the duration time from the calibration com- mand being issued to drdy going low is 9 1/output rate. this is made up of 3 1/output rate for the zero-scale calibra- tion, 3 1/output rate for the full-scale calibration, 3 1/ output rate for a conversion on the analog input and some overhead to set up the coefficients correctly. if drdy is low before (or goes low during) the calibration command write to the setup register, it may take up to one modulator cycle (mclk in/128) before drdy goes high to indicate that cali- bration is in progress. therefore, drdy should be ignored for up to one modulator cycle after the last bit is written to the setup register in the calibration command. for bipolar input ranges in the self-calibrating mode, the se- quence is very similar to that just outlined. in this case, the two points are exactly the same as above, but since the part is config- ured for bipolar operation, the shorted inputs point is actually midscale of the transfer function. system calibration system calibration allows the AD7715 to compensate for system gain and offset errors as well as its own internal errors. system calibration performs the same slope factor calculations as self- calibration but uses voltage values presented by the system to the ain inputs for the zero- and full-scale points. full system calibration requires a two step process, a zs system calibration followed by a fs system calibration. for a full system calibration, the zero-scale point must be pre- sented to the converter first. it must be applied to the converter before the calibration step is initiated and remain stable until the step is complete. once the system zero scale voltage has been set up, a zs system calib ration is then initiated by writing the ap- propriate values (1, 0) to the md1 and md0 bits of the setup register. the zero-scale system calibration is performed at the selected gain. the duration of the calibration is 3 1/output rate. at this time the md1 and md0 bits in the setup register return to 0, 0. this gives the earliest indication that the calibration sequence is complete. the drdy line goes high when calibration is initiated and does not return low until there is a valid new word in the data register. the duration time from the calibra- tion command being issued to drdy going low is 4 1/output rate as the part performs a normal conversion on the ain volt- age before drdy goes low. if drdy is low before (or goes low during) the calibration command write to the setup register, it may take up to one modulator cycle (mclk in/128) before drdy goes high to indicate that calibration is in progress. therefore, drdy should be ignored for up to one modulator cycle after the last bit is written to the setup register in the calibration command. after the zero-scale point is calibrated, the full-scale point is applied to ain and the second step of the calibration process is initiated by again writing the appropriate values (1, 1) to md1 and md0. again the full-scale voltage must be set up before the calibration is initiated and it must remain stable throughout the calibration step. the full-scale system calibration is per- formed at the selected gain. the duration of the calibration is 3 1/output rate. at this time the md1 and md0 bits in the setup register return to 0, 0. this gives the earliest indication that the calibration sequence is complete. the drdy line goes high when calibration is initiated and does not return low until there is a valid new word in the data register. the duration time from the calibration command being issued to drdy going low is 4 1/output rate as the part performs a normal conversion on the ain voltage before drdy goes low. if drdy is low before (or goes low during) the calibration command, write to the setup register, it may take up to one modulator cycle (mclk in/128) before drdy goes high to indicate that cali- bration is in progress. therefore, drdy should be ignored for up to one modulator cycle after the last bit is written to the setup register in the calibration command. in the unipolar mode, the system calibration is performed be- tween the two endpoints of the transfer function; in the bipolar mode, it is performed between midscale (zero differential volt- age) and positive full scale. the fact that the system calibration is a two-step calibration offers another feature. after the sequence of a full system cali- bration has been completed, additional offset or gain calibra- tions can be performed by themselves to adjust the system zero reference point or the system gain. calibrating one of the pa- rameters, either system offset or system gain, will not affect the other parameter. system calibration can also be used to remove any errors from source impedances on the analog input when the part is used in unbuffered mode. a simple r, c antialiasing filter on the front end may introduce a gain error on the analog input voltage but the system calibration can be used to remove this error. span and offset limits whenever a system calibration mode is used, there are limits on the amount of offset and span which can be accommodated. the overriding requirement in determining the amount of offset
AD7715 C19C rev. b and gain that can be accommodated by the part is the require- ment that the positive full-scale calibration limit is 1.05 v ref /gain. this allows the input range to go 5% above the nominal range. the in-built headroom in the AD7715s analog modulator ensures that the part will still operate correctly with a positive full-scale voltage which is 5% beyond the nominal. the range of input span in both the unipolar and bipolar modes has a minimum value of 0.8 v ref /gain and a maximum value of 2.1 v ref /gain. however, the span (which is the difference between the bottom of the AD7715s input range and the top of its input range) must take into account the limitation on the positive full-scale voltage. the amount of offset that can be accommodated depends on whether the unipolar or bipolar mode is being used. once again, the offset must take into ac- count the limitation on the positive full-scale voltage. in unipo- lar mode, there is considerable flexibility in handling negative (with respect to ain(C)) offsets. in both unipolar and bipolar modes, the range of positive offsets which can be handled by the part depends on the selected span. therefore, in determining the limits for system zero-scale and full-scale calibrations, the user has to ensure that the offset range plus the span range does exceed 1.05 v ref /gain. this is best illustrated by looking at a few examples. if the part is used in unipolar mode with a required span of 0.8 v ref /gain, then the offset range which the system cali- bration can handle is from C1.05 v ref /gain to +0.25 v ref / gain. if the part is used in unipolar mode with a required span of v ref /gain, then the offset range which the system calibration can handle is from C1.05 v ref /gain to +0.05 v ref /gain. simi- larly, if the part is used in unipolar mode and required to re- move an offset of 0.2 v ref /gain, then the span range which the system calibration can handle is 0.85 v ref /gain. if the part is used in bipolar mode with a required span of 0.4 v ref /gain, then the offset range which the system cali- bration can handle is from C0.65 v ref /gain to + 0.65 v ref / gain. if the part is used in bipolar mode with a required span of v ref /gain, then the offset range which the system calibra- tion can handle is from C0.05 v ref /gain to +0.05 v ref / gain. similarly, if the part is used in bipolar mode and required to remove an offset of 0.2 v ref /gain, then the span range which the system calibration can handle is 0.85 v ref /gain. power-up and calibration on power-up, the AD7715 performs an internal reset that sets the contents of the internal registers to a known state. there are default values loaded to all registers after a power-on or reset. the default values contain nominal calibration coefficients for the calibration registers. however, to ensure correct calibra- tion for the device a calibration routine should be performed after power-up. the power dissipation and temperature drift of the AD7715 are low, and no warm-up time is required before the initial calibra- tion is performed. however, if an external reference is being used, this reference must have stabilized before calibration is initiated. similarly, if the clock source for the part is generated from a crystal or resonator across the mclk pins, the start-up time for the oscillator circuit should elapse before a calibration is initiated on the part (see below). using the AD7715 clocking and oscillator circuit the AD7715 requires a master clock input, which may be an external cmos compatible clock signal applied to the mclk in pin with the mclk out pin left unconnected. alternatively, a crystal or ceramic resonator of the correct frequency can be connected between mclk in and mclk out in which case the clock circuit will function as an oscillator, providing the clock source for the part. the input sampling frequency, the modulator sampling frequency, the C3 db frequency, output update rate and calibration time are all directly related to the master clock frequency, f clk in . reducing the master clock frequency by a factor of 2 will halve the above frequencies and update rate and double the calibration time. the current drawn from the dv dd power supply is also directly related to f clk in . reducing f clk in by a factor of 2 will halve the dv dd current but will not affect the current drawn from the av dd power supply. using the part with a crystal or ceramic resonator between the mclk in and mclk out pins generally causes more cur- rent to be drawn from dv dd than when the part is clocked from a driven clock signal at the mclk in pin. this is because the on-chip oscillator circuit is active in the case of the crystal or ceramic resonator. therefore, the lowest possible current on the AD7715 is achieved with an externally applied clock at the mclk in pin with mclk out unconnected and unloaded. the amount of additional current taken by the oscillator de- pends on a number of factorsfirst, the larger the value of capacitor placed on the mclk in and mclk out pins, then the larger the dv dd current consumption on the AD7715. care should be taken not to exceed the capacitor values recommended by the crystal and ceramic resonator manufacturers to avoid consuming unnecessary dv dd current. typical values recom- mended by crystal or ceramic resonator manufacturers are in the range of 30 pf to 50 pf, and if the capacitor values on mclk in and mclk out are kept in this range, they will not result in any excessive dv dd current. another factor that influences the dv dd current is the effective series resistance (esr) of the crystal which appears between the mclk in and mclk out pins of the AD7715. as a general rule, the lower the esr value then the lower the current taken by the oscillator circuit. when operating with a clock frequency of 2.4576 mhz, there is 50 m a difference in the dv dd current between an externally applied clock and a crystal resonator when operating with a dv dd of +3 v. with dv dd = +5 v and f clk in = 2.4576 mhz, the typical dv dd current increases by 200 m a for a crystal/ resonator supplied clock versus an externally applied clock. the esr values for crystals and resonators at this frequency tend to be low and as a result there tends to be little difference between different crystal and resonator types. when operating with a clock frequency of 1 mhz, the esr value for different crystal types varies significantly. as a result, the dv dd current drain varies across crystal types. when using a crystal with an esr of 700 w or when using a ceramic resonator, the increase in the typical dv dd current over an externally-applied clock is 50 m a with dv dd = +3 v and 175 m a with dv dd = +5 v. when using a crystal with an esr of 3 k w , the increase in the typical dv dd current over an externally applied clock is 100 m a with dv dd = +3 v and 400 m a with dv dd = +5 v.
rev. b AD7715 C20C the on-chip oscillator circuit also has a start-up time associated with it before it is oscillating at its correct frequency and correct voltage levels. the typical start-up time for the circuit is 10 ms with a dv dd of +5 v and 15 ms with a dv dd of +3 v. at 3 v supplies, depending on the loading capacitances on the mclk pins, a 1 m w feedback resistor may be required across the crys- tal or resonator in order to keep the start up times around the 15 ms duration. the AD7715s master clock appears on the mclk out pin of the device. the maximum recommended load on this pin is one cmos load. when using a crystal or ceramic resonator to gen- erate the AD7715s clock, it may be desirable to then use this clock as the clock source for the system. in this case, it is recom- mended that the mclk out signal is buffered with a cmos buffer before being applied to the rest of the circuit. system synchronization the fsync bit of the setup register allows the user to reset the modulator and digital filter without affecting any of the setup conditions on the part. this allows the user to start gath- ering samples of the analog input from a known point in time, i.e., when the fsync is changed from 1 to 0. with a 1 in the fsync bit of the setup register, the digital filter and analog modulator are held in a known reset state and the part is not processing any input samples. when a 0 is then written to the fsync bit, the modulator and filter are taken out of this reset state and on the next master clock edge the part starts to gather samples again. the fsync input can also be used as a software start convert command allowing the AD7715 to be operated in a conven- tional converter fashion. in this mode, writing to the fsync bit starts conversion and the falling edge of drdy indicates when conversion is complete. the disadvantage of this scheme is that the settling time of the filter has to be taken into account for every data register update. this means that the rate at which the data register is updated is three times slower in this mode. since the fsync bit resets the digital filter, the full settling time of 3 1/output rate must elapse before there is a new word loaded to the output register on the part. if the drdy signal is low when fsync goes to a 0, the drdy signal will not be reset high by the fsync command. this is because the AD7715 recognizes that there is a word in the data register that has not been read. the drdy line will stay low until an update of the data register takes place at which time it will go high for 500 t clk in before returning low again. a read from the data register resets the drdy signal high, and it will not return low until the settling time of the filter has elapsed (from the fsync command) and there is a valid new word in the data register. if the drdy line is high when the fsync command is issued, the drdy line will not return low until the settling time of the filter has elapsed. reset input the reset input on the AD7715 resets all the logic, the digital filter and the analog modulator while all on-chip registers are reset to their default state. drdy is driven high and the AD7715 ignores all communications to any of its registers while the reset input is low. when the reset input returns high, the AD7715 starts to process data, and drdy will return low in 3 1/output rate indicating a valid new word in the data register. however, the AD7715 operates with its default setup cond itions after a reset and it is generally necessary to set up all registers and carry out a calibration after a reset command. the AD7715s on-chip oscillator circuit continues to function even when the reset input is low. the master clock signal continues to be available on the mclk out pin. therefore, in applications w here the system clock is provided by the AD7715s clock, the AD7715 produces an uninterrupted master clock during reset commands. standby mode the stby bit in the communications register of the AD7715 allows the user to place the part in a power-down mode when it is not required to provide conversion results. the AD7715 retains the contents of all its on-chip registers (including the data register) while in standby mode. when released from standby mode, the part starts to process data and a new word is available in the data register in 3 1/output rate from when a 0 is written to the stby bit. the stby bit does not affect the digital interface, and it does not affect the status of the drdy line. if drdy is high when the stby bit is brought low, it will remain high until there is a valid new word in the data register. if drdy is low when the stby bit is brought low, it will remain low until the data regis- ter is updated at which time the drdy line will return high for 500 t clk in before returning low again. if drdy is low when the part enters its standby mode (indicating a valid unread word in the data register), the data register can be read while the part is in standby. at the end of this read operation, the drdy will be reset high as normal. placing the part in standby mode reduces the total current to 5 m a typical when the part is operated from an external master clock provided this master clock is stopped. if the external clock continues to run in standby mode, the standby current increases to 150 m a typical with 5 v supplies and 75 m a typical with 3.3 v supplies. if a crystal or ceramic resonator is used as the clock source, then the total current in standby mode is 400 m a typical with 5 v supplies and 90 m a with 3.3 v supplies. this is because the on-chip oscillator circuit continues to run when the part is in its standby mode. this is important in applications where the system clock is provided by the AD7715s clock, so that the AD7715 produces an uninterrupted master clock even when it is in its standby mode. accuracy sigma-delta adcs, like vfcs and other integrating adcs, do not contain any source of nonmonotonicity and inherently offer no missing codes performance. the AD7715 achieves excellent linearity by the use of high quality, on-chip capacitors, which have a very low capacitance/voltage coefficient. the device also achieves low input drift through the use of chopper-stabilized techniques in its input stage. to ensure excellent performance over time and temperature, the AD7715 uses digital calibration techniques which minimize offset and gain error. drift considerations the AD7715 uses chopper stabilization techniques to minimize input offset drift. charge injection in the analog switches and dc leakage currents at the sampling node are the primary sources of offset voltage drift in the converter. the dc input leakage current is essentially independent of the selected gain. gain drift within the converter depends primarily upon the temperature tracking of the internal capacitors. it is not af- fected by leakage currents.
AD7715 C21C rev. b measurement errors due to offset drift or gain drift can be elimi- nated at any time by recalibrating the converter. using the sys- tem calibration mode can also minimize offset and gain errors in the signal conditioning circuitry. integral and differential l inear- ity errors are not significantly affected by temperature changes. power supplies there is no specific power sequence required for the AD7715; either the av dd or the dv dd supply can come up first. while the latch-up performance of the AD7715 is good, it is important that power is applied to the AD7715 before signals at ref in, ain or the logic input pins in order to avoid excessive currents. if this is not possible, then the current which flows in any of these pins should be limited. if separate supplies are used for the AD7715 and the system digital circuitry, then the AD7715 should be powered up first. if it is not possible to guarantee this, then current limiting resistors should be placed in series with the logic inputs to again limit the current. supply current the current consumption on the AD7715 is specified for sup- plies in the range +3 v to +3.6 v and in the range +4.75 v to +5.25 v. the part operates over a +2.85 v to +5.25 v supply range and the i dd for the part varies as the supply voltage varies over this range. figure 5 shows the variation of the typical i dd with v dd voltage for both a 1 mhz external clock and a 2.4576 mhz external clock at +25 c. the AD7715 is operated in unbuffered mode. the relationship shows that the i dd is minimized by operating the part with lower v dd voltages. i dd on the AD7715 is also minimized by using an external master clock or by optimizing external components when using the on- chip oscillator circuit. supply voltage (av dd & dv dd ) C volts 0 2.85 0.9 1.0 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 3.15 3.45 4.05 4.35 4.65 4.95 5.25 3.75 mclk in = 2.4576mhz mclk in = 1mhz supply current (av dd & dv dd ) C ma figure 5. i dd vs. supply voltage grounding and layout since the analog inputs and reference input are differential, most of the voltages in the analog modulator are common-mode voltages. the excellent common-mode rejection of the part will remove common-mode noise on these inputs. the analog and digital supplies to the AD7715 are independent and separately pinned out to minimize coupling between the analog and digital sections of the device. the digital filter will provide rejection of broadband noise on the power supplies, except at integer mul- tiples of the modulator sampling frequency. the digital filter also removes noise from the analog and reference inputs pro- vided those noise sources do not saturate the analog modulator. as a result, the AD7715 is more immune to noise interference that a conventional high resolution converter. however, because the resolution of the AD7715 is so high and the noise levels from the AD7715 so low, care must be taken with regard to grounding and layout. the printed circuit board which houses the AD7715 should be designed such that the analog and digital sections are separated and confined to certain areas of the board. this facilitates the use of ground planes which can be separated easily. a minimum etch technique is generally best for ground planes as it gives the best shielding. digital and analog ground planes should only be joined in one place. if the AD7715 is the only device requiring an agnd to dgnd connection, then the ground planes should be connected at the agnd and dgnd pins of the AD7715. if the AD7715 is in a system where multiple devices require agnd to dgnd connections, the connection should still be made at one point only, a star ground point which should be established as close as possible to the AD7715. avoid running digital lines under the device as these will couple noise onto the die. the analog ground plane should be allowed to run under the AD7715 to avoid noise coupling. the power supply lines to the AD7715 should use as large a trace as pos- sible to provide low impedance paths and reduce the effects of glitches on the power supply line. fast switching signals like clocks should be shielded with digital ground to avoid radiating noise to other sections of the board and clock signals should never be run near the analog inputs. avoid crossover of digital and analog signals. traces on opposite sides of the board should run at right angles to each other. this will reduce the effects of feedthrough through the board. a microstrip technique is by far the best but is not always possible with a double-sided board. in this technique, the component side of the board is dedicated to ground planes while signals are placed on the solder side. good decoupling is important when using high resolution adcs. all analog supplies should be decoupled with 10 m f tantalum in parallel with 0.1 m f capacitors to agnd. to achieve the best from these decoupling components, they must be placed as close as possible to the device, ideally right up against the device. all logic chips should be decoupled with 0.1 m f disc ceramic capacitors to dgnd. in systems where a common supply voltage is used to drive both the av dd and dv dd of the AD7715, it is recommended that the systems av dd supply is used. this supply should have the recom- mended analog supply decoupling capacitors between the av dd pin of the AD7715 and agnd and the recommended digital supply decoupling capacitor between the dv dd pin of the AD7715 and dgnd. evaluating the AD7715 performance the recommended layout for the AD7715 is outlined in the evaluation board for the AD7715. the evaluation board pack- age includes a fully assembled and tested evaluation board, documentation, software for controlling the board over the printer port of a pc and software for analyzing the AD7715s performance on the pc. for the AD7715-5, the evaluation board order number is eval-AD7715-5eb and for the AD7715-3, the order number is eval-AD7715-3eb. noise levels in the signals applied to the AD7715 may also affect performance of the part. the AD7715 software evaluation package allows the user to evaluate the true performance of the part, independent of the analog input signal. the scheme
rev. b AD7715 C22C involves using a test mode on the part where the differential inputs to the AD7715 are internally shorted together to provide a zero differential voltage for the analog modulator. external to the device, the ain(C) input should be connected to a voltage which is within the allowable common-mode range of the part. this scheme should be used after a calibration has been per- formed on the part. digital interface the AD7715s programmable functions are controlled using a set of on-chip registers as outlined previously. data is written to these registers via the parts serial interface and read access to the on-chip registers is also provided by this interface. all com- munications to the part must start with a write operation to the communications register. after power-on or reset, the de- vice expects a write to its communications register. the data written to this register determines whether the next operation to the part is a read or a write operation and also determines to which register this read or write operation occurs. therefore, write access to any of the other registers on the part starts with a write operation to the communications register followed by a write to the selected register. a read operation from any other register on the part (including the output data register) starts with a write operation to the communications register followed by a read operation from the selected register. the AD7715s serial interface consists of five signals, cs , sclk, din, dout and drdy . the din line is used for transferring data into the on-chip registers while the dout line is used for accessing data from the on-chip registers. sclk is the serial clock input for the device and all data transfers (either on din or dout) take place with respect to this sclk signal. the drdy line is used as a status signal to indicate when data is ready to be read from the AD7715s data register. drdy goes low when a new data word is available in the output regis- ter. it is reset high when a read operation from the data register is complete. it also goes high prior to the updating of the output register to indicate when not to read from the device to ensure that a data read is not attempted while the register is being updated. cs is used to select the device. it can be used to de- code the AD7715 in systems where a number of parts are con- nected to the serial bus. figures 5 and 6 show timing diagrams for interfacing to the AD7715 with cs used to decode the part. figure 5 is for a read operation from the AD7715s output shift register, while figure 6 shows a write operation to the input shift register. it is pos- sible to read the same data twice from the output register even though the drdy line returns high after the first read opera- tion. care must be taken, however, to ensure that the read operations have been completed before the next output update is about to take place. the AD7715 serial interface can operate in three-wire mode by tying the cs input low. in this case, the sclk, din and dout lines are used to communicate with the AD7715 and the status of drdy can be obtained by interrogating the msb of the communications register. this scheme is suitable for interfacing to microcontrollers. if cs is required as a decoding signal, it can be generated from a port bit. for microcontroller interfaces, it is recommended that the sclk idles high between data transfers. the AD7715 can also be operated with cs used as a frame synchronization signal. this scheme is suitable for dsp inter- faces. in this case, the first bit (msb) is effectively clocked out by cs since cs would normally occur after the falling edge of sclk in dsps. the sclk can continue to run between data transfers provided the timing numbers are obeyed. the serial interface can be reset by exercising the reset input on the part. it can also be reset by writing a series of 1s on the din input. if a logic 1 is written to the AD7715 din line for at least 32 serial clock cycles, the serial interface is reset. this ensures that in three-wire systems that if the interface gets lost either via a software error or by some glitch in the system, it can be reset back into a known state. this state returns the interface dout sclk cs drdy msb t 5 t 7 t 9 lsb t 8 t 6 t 3 t 10 t 4 figure 6. read cycle timing diagram din sclk cs msb t 12 t 15 lsb t 16 t 14 t 13 t 11 figure 7. write cycle timing diagram
AD7715 C23C rev. b to where the AD7715 is expecting a write operation to its com- munications register. this operation in itself does not reset the contents of any registers, but since the interface was lost, the information which was written to any of the registers is un- known and it is advisable to set up all registers again. some microprocessor or microcontroller serial interfaces have a single serial data line. in this case, it is possible to connect the AD7715s data out and data in lines together and con- nect then to the single data line of the processor. a 10 k w pull- up resistor should be used on this single data line. in this case, if the interface gets lost, because the read and write operations share the same line the procedure to reset it back to a known state is somewhat different than described previously. it requires a read operation of 24 serial clocks followed by a write operation where a logic 1 is written for at least 32 serial clock cycles to ensure that the serial interface is back into a known state. configuring the AD7715 the AD7715 contains three on-chip registers which the user accesses via the serial interface. communication with any of these registers is initiated by writing to the communications register first. figure 8 outlines a flow diagram of the sequence which is used to configure all registers after a power-up or reset. the flowchart also shows two different read optionsthe first where the drdy pin is polled to determine when an update of the data register has taken place, the second where the drdy bit of the communications register is interrogated to see if a data register update has taken place. also included in the flow- ing diagram is a series of words which should be written to the registers for a particular set of operating conditions. these con- ditions are gain of 1, no filter sync, bipolar mode, buffer off, clock of 2.4576 mhz and an output rate of 60 hz. start configure & initialize m c/ m p serial port power-on/reset for AD7715 write to communications register setting up gain & setting up next operation to be a write to the setup register (10 hex) write to setup register setting up required values & initiating a self calibration (68 hex) poll drdy pin drdy low? yes no write to communications register setting up same gain & setting up next operation to be a read from the data register (38 hex) read from data register poll drdy bit of communications register yes no write to communications register setting up same gain & setting up next operation to be a read from the data register (38 hex) write to communications register setting up same gain & setting up next operation to be a read from the communications register (08 hex) read from communications register drdy low? read from data register figure 8. flowchart for setting up and reading from the AD7715
rev. b AD7715 C24C microcomputer/microprocessor interfacing the AD7715s flexible serial interface allows for easy interface to most microcomputers and microprocessors. the flowchart of figure 8 outlines the sequence which should be followed when interfacing a microcontroller or microprocessor to the AD7715. figures 9, 10 and 11 show some typical interface circuits. the serial interface on the AD7715 has the capability of operat- ing from just three wires and is compatible with spi interface protocols. the three-wire operation makes the part ideal for isolated systems where minimizing the number of interface lines minimizes the number of opto-isolators required in the system. the rise and fall times of the digital inputs to the AD7715 (especially the sclk input) should be no longer than 1 m s. most of the registers on the AD7715 are 8-bit registers. this facilitates easy interfacing to the 8-bit serial ports of microcon- trollers. some of the registers on the part are up to 16 bits, but data transfers to these 16-bit registers can consist of a full 16-bit transfer or two 8-bit transfers to the serial port of the microcon- troller. dsp processors and microprocessors generally transfer 16 bits of data in a serial data operation. some of these proces- sors, such as the adsp-2105, have the facility to program the amount of cycles in a serial transfer. this allows the user to tailor the number of bits in any transfer to match the register length of the required register in the AD7715. even though some of the registers on the AD7715 are only eight bits in length, communicating with two of these registers in successive write operations can be handled as a single 16-bit data transfer if required. for example, if the setup register is to be updated, the processor must first write to the communica- tions register (saying that the next operation is a write to the setup register) and then write eight bits to the setup register. this can all be done in a single 16-bit transfer if required be- cause once the eight serial clocks of the write operation to the communications register have been completed, the part imme- diately sets itself up for a write operation to the setup register. AD7715 to 68hc11 interface figure 9 shows an interface between the AD7715 and the 68hc11 microcontroller. the diagram shows the minimum (three-wire) interface with cs on the AD7715 hardwired low. in this scheme, the drdy bit of the communications register is monitored to determine when the data register is updated. an alternative scheme, which increases the number of interface AD7715 data out sclk cs 68hc11 ss dv dd reset data in sck miso mosi dv dd figure 9. AD7715 to 68hc11 interface lines to four, is to monitor the drdy output line from the AD7715. the monitoring of the drdy line can be done in two ways. first, drdy can be connected to one of the 68hc11s port bits (such as pc0) which is configured as an input. this port bit is then polled to determine the status of drdy . the second scheme is to use an interrupt driven system, in which case the drdy output is connected to the irq input of the 68hc11. for interfaces that require control of the cs input on the AD7715, one of the port bits of the 68hc11 (such as pc1), which is configured as an output, can be used to drive the cs input. the 68hc11 is configured in the master mode with its cpol bit set to a logic one and its cpha bit set to a logic one. when the 68hc11 is configured like this, its sclk line idles high between data transfers. the AD7715 is not capable of full du- plex operation. if the AD7715 is configured for a write opera- tion, no data appears on the data out lines even when the sclk input is active. similarly, if the AD7715 is configured for a read operation, data presented to the part on the data in line is ignored even when sclk is active. coding for an interface between the 68hc11 and the AD7715 is given in table xvi. in this example, the drdy output line of the AD7715 is connected to the pc0 port bit of the 68hc11 and is polled to determine its status.
AD7715 C25C rev. b AD7715 to 8xc51 interface an interface circuit between the AD7715 and the 8xc51 microcontroller is shown in figure 10. the diagram shows the minimum number of interface connections with cs on the AD7715 hardwired low. in the case of the 8xc51 interface, the minimum number of interconnects is just two. in this scheme, the drdy bit of the communications register is monitored to determine when the data register is updated. the alternative scheme, which increases the number of interface lines to three, is to monitor the drdy output line from the AD7715. the monitoring of the drdy line can be done in two ways. first, drdy can be connected to one of the 8xc51s port bits (such as p1.0) which is configured as an input. this port bit is then polled to determine the status of drdy . the second scheme is to use an interrupt driven system in which case, the drdy output is connected to the int1 input of the 8xc51. for inter- faces that require control of the cs input on the AD7715, one of the port bits of the 8xc51 (such as p1.1), which is config- ured as an output, can be used to drive the cs input. the 8xc51 is configured in its mode 0 serial interface mode. its serial interface contains a single data line. as a result, the data out and data in pins of the AD7715 should be connected together with a 10 k w pull-up resistor. the serial clock on the 8xc51 idles high between data transfers. the 8xc51 outputs the lsb first in a write operation while the AD7715 rearranged before being written to the output serial register. similarly, the AD7715 outputs the msb first during a read operation while the 8xc51 expects the lsb first. there- fore, the data which is read into the serial buffer needs to be rearranged before the correct data word from the AD7715 is available in the accumulator. AD7715 data out cs 8xc51 reset data in sclk p3.0 p3.1 dv dd dv dd 10k v figure 10. AD7715 to 8xc51 interface AD7715 to adsp-2103/adsp-2105 interface figure 11 shows an interface between the AD7715 and the adsp-2103/adsp-2105 dsp processor. in the interface shown, the drdy bit of the communications register is again monitored to determine when the data register is updated. the alternative scheme is to use an interrupt driven system, in which case the drdy output is connected to the irq2 input of the adsp-2103/adsp-2105. the serial interface of the adsp- 2103/adsp-2105 is set up for alternate framing mode. the rfs and tfs pins of the adsp-2103/adsp-2105 are config- ured as active low outputs, and the adsp-2103/adsp-2105 serial clock line, sclk, is also configured as an output. the cs for the AD7715 is active when either the rfs or tfs outputs from the adsp-2103/adsp-2105 are active. the serial clock rate on the adsp-2103/adsp-2105 should be limited to 3 mhz to ensure correct operation with the AD7715. AD7715 data out cs adsp-2103/2105 reset data in sclk rfs dr dt dv dd tfs sclk figure 11. AD7715 to adsp-2103/adsp-2105 interface code for setting up the AD7715 table xvi gives a set of read and write routines in c code for interfacing the 68hc11 microcontroller to the AD7715. the sample program sets up the various registers on the AD7715 and reads 1000 samples from the part into the 68hc11. the setup conditions on the part are exactly the same as those out- lined for the flowchart of figure 8. in the example code given here, the drdy output is polled to determine if a new valid word is available in the data register. the sequence of the events in this program are as follows: 1. write to the communications register, setting the gain to 1 with standby inactive. 2. write to the setup register, setting bipolar mode, buffer off, no filter synchronization, confirming a clock frequency of 2.4576 mhz, setting the output rate for 60 hz and initiating a self-calibration. 3. poll the drdy output. 4. read the data from the data register. 5. loop around doing steps 3 and 4 until the specified number of samples have been taken.
rev. b AD7715 C26C table xvi. c code for interfacing AD7715 to 68hc11 /* this program has read and write routines for the 68hc11 to interface to the AD7715 and the sample program sets the various registers and then reads 1000 samples from the part. */ #include #include #define num_samples 1000 /* change the number of data samples */ #define max_reg_length 2 /* this says that the max length of a register is 2 bytes */ writetoreg (int); read (int,char); char *datapointer = store; char store[num_samples*max_reg_length + 30]; void main() { /* the only pin that is programmed here from the 68hc11 is the /cs and this is why the pc2 bit of portc is made as an output */ char a; ddrc = 0x04; /* pc2 is an output the rest of the port bits are inputs */ portc | = 0x04; /* make the /cs line high */ writetoreg(0x10); /* set the gain to 1, standby off and set the next operation as write to the setup register */ writetoreg(0x68); /* set bipolar mode, buffer off, no filter sync, confirm clock as 2.4576mhz, set output rate to 60hz and do a self calibration */ while(portc & 0x10); /* wait for /drdy to go low */ for(a=0;a AD7715 C27C rev. b +20 mv to 0 v to +2.5 v and bipolar inputs of 20 mv to 2.5 v. because the part operates from a single supply, these bipolar ranges are with respect to a biased-up differential input. pressure measurement one typical application of the AD7715 is pressure measurement. figure 12 shows the AD7715 used with a pressure transducer, the bp01 from sensym. the pressure transducer is arranged in a bridge network and gives a differential output voltage between its out(+) and out(C) terminals. with rated full-scale pres- sure (in this case 300 mmhg) on the transducer, the differential output voltage is 3 mv/v of the input voltage (i.e., the voltage between its in(+) and in(C) terminals). assuming a 5 v excitation voltage, the full-scale output range from the transducer is 15 mv. the excitation voltage for the bridge is also used to generate the reference voltage for the AD7715. therefore, variations in the excitation voltage do not introduce errors in the system. choosing resistor values of 24 k w and 15 k w as per the diagram give a 1.92 v reference voltage for the AD7715 when the excitation voltage is 5 v. using the part with a programmed gain of 128 results in the full- scale input span of the AD7715 being 15 mv which corresponds with the output span from the transducer. applications the AD7715 provides a low cost, high resolution analog-to- digital function. because the analog-to-digital function is pro- vided by a sigma-delta architecture, it makes the part more immune to noisy environments thus making the part ideal for use in industrial and process control applications. it also provides a programmable gain amplifier, a digital filter and calibration options. thus, it provides far more system level functionality than off-the-shelf integrating adcs without the disadvantage of having to supply a high quality integrating ca- pacitor. in addition, using the AD7715 in a system allows the system designer to achieve a much higher level of resolution because noise performance of the AD7715 is significantly better than that of the integrating adcs. the on-chip pga allows the AD7715 to handle an analog input voltage range as low as 10 mv full-scale with v ref = +1.25 v. the differential inputs of the part allow this analog input range to have an absolute value anywhere between agnd and av dd when the part is operated in unbuffered mode. it allows the user to connect the transducer directly to the input of the AD7715. the programmable gain front end on the AD7715 allows the part to handle unipolar analog input ranges from 0 mv to inC out+ in+ outC auto-zeroed modulator charge balancing a/d converter digital filter av dd serial interface register bank mclk in mclk out ref in (+) ref in (C) agnd dgnd dout din cs sclk drdy reset 24k v 15k v dv dd AD7715 +5v excitation voltage = +5v ain(+) ain(C) a = 1C128 pga buffer clock generation figure 12. pressure measurement using the AD7715
rev. b AD7715 C28C temperature measurement another application area for the AD7715 is in temperature measurement. figure 13 outlines a connection from a thermo- couple to the AD7715. in this application, the AD7715 is oper- ated in its buffered mode to allow large decoupling capacitors on the front end to eliminate any noise pickup which there may have been in the thermocouple leads. when the AD7715 is operated in buffered mode, it has a reduced common-mode range. in order to place the differential voltage from the thermo- couple on a suitable common-mode voltage, the ain(C) input of the AD7715 is biased up at the reference voltage, +2.5 v. figure 14 shows another temperature measurement application for the AD7715. in this case, the transducer is an rtd (resis- tive temperature device), a pt100. the arrangement is a 4- lead rtd configuration. there are voltage drops across the lead resistances r l1 and r l4 , but these simply shift the common- mode voltage. there is no voltage drop across lead resistances r l2 and r l3 as the input current to the AD7715 is very low . the lead resistances present a small source impedance so it would not generally be necessary to turn on the buffer on the AD7715. if the buffer is required, the common-mode voltage should be set accordingly by inserting a small resistance between the bot- tom end of the rtd and agnd of the AD7715. in the appli- cation shown an external 400 m a current source provides the excitation current for the pt100 and it also generates the refer- ence voltage for the AD7715 via the 6.25 k w resistor. variations in the excitation current do not affect the circuit as both the input voltage and the reference voltage vary ratiometrically with the excitation current. however, the 6.25 k w resistor must have a low temperature coefficient to avoid errors in the reference voltage over temperature. auto-zeroed modulator charge balancing a/d converter digital filter av dd serial interface register bank mclk in mclk out ref in (+) ref in (C) agnd dgnd dout din cs sclk drdy reset dv dd AD7715 +5v ain (+) ain (C) a = 1C128 pga buffer clock generation gnd cc v out ref192 thermocouple junction +5v +v in r r figure 13. thermocouple measurement using the AD7715 400 m a r l4 r l3 r l2 r l1 rtd 6.25k v auto-zeroed modulator charge balancing a/d converter digital filter av dd serial interface register bank mclk in mclk out ref in (+) ref in (C) agnd dgnd dout din cs sclk drdy reset dv dd AD7715 +5v ain(+) ain(C) a = 1C128 pga buffer clock generation figure 14. rtd measurement using the AD7715
AD7715 C29C rev. b smart transmitters another area where the low power, single supply, three-wire interface capabilities is of benefit is in smart transmitters. here, the entire smart transmitter must operate from the 4 ma to 20 ma loop. tolerances in the loop mean that the amount of current available to power the transmitter is as low as 3.5 ma. the AD7715 consumes only 450 m a, leaving 3 ma available for the rest of the transmitter. figure 15 shows a block diagram of a smart transmitter which includes the AD7715. not shown in figure 15 is the isolated power source required to power the front end. input/output stage signal conditioner microcontroller unit *pid *range setting *calibration *linearization *output control *serial communication *hart protocol voltage reference mclk out AD7715 isolated supply isolation barrier sensors rtd mv ohm tc loop rtn mclk in com 3v 4C20ma dv dd av dd ref in agnd dgnd isolated ground com v cc 3v main transmitter assembly bandpass filter waveform shaper hart modem bell 202 d/a converter voltage reference voltage regulator figure 15. smart transmitter using the AD7715
rev. b AD7715 C30C page index topic page features . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 general description . . . . . . . . . . . . . . . . . . . . . . . . . 1 product highlights . . . . . . . . . . . . . . . . . . . . . . . . . 1 AD7715-5 specifications . . . . . . . . . . . . . . . . . . . . . . . 2 AD7715-3 specifications . . . . . . . . . . . . . . . . . . . . . . . 3 specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 timing characteristics . . . . . . . . . . . . . . . . . . . . . 5 absolute maximum ratings . . . . . . . . . . . . . . . . . 6 ordering guide . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 pin configurations . . . . . . . . . . . . . . . . . . . . . . . . . . 6 pin function description . . . . . . . . . . . . . . . . . . . 7 terminology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 on-chip registers . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 communications register . . . . . . . . . . . . . . . . . . . . . . . . . 9 setup register . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 test register . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 data register . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 output noise . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 AD7715-5 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 AD7715-3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 AD7715-3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 calibration sequences . . . . . . . . . . . . . . . . . . . . . 14 circuit description . . . . . . . . . . . . . . . . . . . . . . . . 14 analog input . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 analog input ranges . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 input sample rate . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 bipolar/unipolar inputs . . . . . . . . . . . . . . . . . . . . . . . . . . 15 reference input . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 digital filtering . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 filter characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 post-filtering . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 analog filtering . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 calibration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 self-calibration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 system calibration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 span and offset limits . . . . . . . . . . . . . . . . . . . . . . . . . . 18 power-up and calibration . . . . . . . . . . . . . . . . . . . . . . . . 19 using the AD7715 . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 clocking and oscillator circuit . . . . . . . . . . . . . . . . . . . . 19 system synchronization . . . . . . . . . . . . . . . . . . . . . . . . . . 20 reset input . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 standby mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 accuracy . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 drift considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 power supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 supply current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 grounding and layout . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 evaluating the AD7715 performance . . . . . . . . . . . . . . . . 21 digital interface . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 configuring the AD7715 . . . . . . . . . . . . . . . . . . . . . 23 microcomputer/microprocessor interfacing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 AD7715 to 68hc11 interface . . . . . . . . . . . . . . . . . . . . . 24 AD7715 to 8xc51 interface . . . . . . . . . . . . . . . . . . . . . . 25 AD7715 to adsp-2103/adsp-2105 interface . . . . . . . . 25 topic page code for setting up AD7715 . . . . . . . . . . . . . . . . . 25 applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27 pressure measurement . . . . . . . . . . . . . . . . . . . . . . . . . . . 27 temperature measurement . . . . . . . . . . . . . . . . . . . . . . . 28 smart transmitters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29 index . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 outline dimensions . . . . . . . . . . . . . . . . . . . . . . . . . 31 table index table title table i communications register . . . . . . . . . . . . . . . . . . 9 table ii register selection . . . . . . . . . . . . . . . . . . . . . . . . 9 table iii setup register . . . . . . . . . . . . . . . . . . . . . . . . . . 10 table iv output update rates . . . . . . . . . . . . . . . . . . . . 11 table v output rms noise vs. gain and output update rate for AD7715-5 (unbuffered mode) . . . . . . 12 table vi peak-to-peak resolution vs. gain and output update rate for AD7715-5 (unbuffered mode) . . 12 table vii o utput rms noise vs. gain and output update rate for AD7715-5 (buffered mode) . . . . . . . . . 12 table viii peak-to-peak resolution vs. gain and output update rate for AD7715-5 (buffered mode) . . 12 table ix output rms noise vs. gain and output update rate for AD7715-3 (unbuffered mode) . . . . . . 13 table x peak-to-peak resolution vs. gain and output update rate for a d7715-3 (unbuffered mode) . . 13 table xi output rms noise vs. gain and output update rate for AD7715-5 (buffered mode) . . . . . . . . 13 table xii peak-to-peak resolution vs. gain and output update rate for AD7715-3 (buffered mode) . . 13 table xiii calibration sequences . . . . . . . . . . . . . . . . . . . . 14 table xiv external r, c combination for no 16-bit gain error . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 table xv input sampling frequency vs. gain . . . . . . . . . 15 table xvi c code for interfacing AD7715 to 68hc11 . . . 26
AD7715 C31C rev. b outline dimensions dimensions shown in inches and (mm). 16-lead plastic dip (n-16) 16 18 9 0.840 (21.33) 0.745 (18.93) 0.280 (7.11) 0.240 (6.10) pin 1 seating plane 0.022 (0.558) 0.014 (0.356) 0.060 (1.52) 0.015 (0.38) 0.210 (5.33) max 0.130 (3.30) min 0.070 (1.77) 0.045 (1.15) 0.100 (2.54) bsc 0.160 (4.06) 0.115 (2.93) 0.325 (8.25) 0.300 (7.62) 0.015 (0.381) 0.008 (0.204) 0.195 (4.95) 0.115 (2.93) 16-lead soic (r-16) 16 9 8 1 0.4133 (10.50) 0.3977 (10.00) 0.4193 (10.65) 0.3937 (10.00) 0.2992 (7.60) 0.2914 (7.40) pin 1 seating plane 0.0118 (0.30) 0.0040 (0.10) 0.0192 (0.49) 0.0138 (0.35) 0.1043 (2.65) 0.0926 (2.35) 0.0500 (1.27) bsc 0.0125 (0.32) 0.0091 (0.23) 0.0500 (1.27) 0.0157 (0.40) 8 0 0.0291 (0.74) 0.0098 (0.25) 3 45 16-lead tssop (ru-16) 16 9 8 1 0.201 (5.10) 0.193 (4.90) 0.256 (6.50) 0.246 (6.25) 0.177 (4.50) 0.169 (4.30) pin 1 seating plane 0.006 (0.15) 0.002 (0.05) 0.0118 (0.30) 0.0075 (0.19) 0.0256 (0.65) bsc 0.0433 (1.10) max 0.0079 (0.20) 0.0035 (0.090) 0.028 (0.70) 0.020 (0.50) 8 0 c2016aC0C7/98 printed in u.s.a.

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